Method for adjusting acquisition speed in a wireless network

ABSTRACT

A method is provided for acquiring incoming signals in a wireless network device. This method uses three different types of preamble: a normal preamble, a short preamble, and a long preamble. One of these preambles will be used as a default preamble. Then, depending upon signal parameters, the device can change from one preamble to another, trading off data transmission speed and acquisition time to achieve the maximum data transmission speed by using the minimum acquisition time. These signal parameters could be signal strength, the number of packet retransmissions the device must request, or any other metric that is required. And thresholds will vary with the quality of service.

CROSS-REFERENCE TO RELATED PATENT DOCUMENTS

This application is a continuation-in-part of U.S. application Ser. No.10/367,834, filed Feb. 19, 2003, entitled “M-ARY ORTHAGONAL CODEDCOMMUNICATIONS METHOD AND SYSTEM,” which relies for priority on U.S.provisional application Ser. No. 60/357,638, by Matthew L. Welborn,filed Feb. 20, 2002, entitled “M-ARY BI-ORTHAGONAL CODED ULTRAWIDEBANDCOMMUNICATIONS SYSTEM,” the contents of each of which are herebyincorporated by reference in their entirety. This application alsorelies for priority on U.S. provisional application Ser. No. 60/397,105,by Matthew L. Welborn et al., filed Jul. 22, 2002, entitled “M-ARYBIORTHAGONAL KEY BINARY PHASE SHIFT KEY SCHEME FOR ULTRAWIDE BANDWIDTHCOMMUNICATIONS USING RANDOM OVERLAY CODES AND FREQUENCY OFFSET FORPICONET SEPARATION,” U.S. provisional application Ser. No. 60/397,104,by Richard D. Roberts, filed Jul. 22, 2002, entitled “METHOD ANDAPPARATUS FOR CARRIER DETECTION FOR CODE DIVISION MULTIPLE ACCESSULTRAWIDE BANDWIDTH COMMUNICATIONS,” and U.S. provisional applicationSer. No. 60/398,596, by Richard D. Roberts, filed Jul. 26, 2002,entitled “METHOD AND SYSTEM OF ACQUIRING A BINARY PHASE SHIFT KEYULTRAWIDE BANDWIDTH SIGNAL,” the contents of all of which are herebyincorporated by reference in their entirety.

BACKGROUND OF THE INVENTION

The present invention relates to ultrawide bandwidth (UWB) transmitters,receivers and transmission schemes. More particularly, the presentinvention relates to a method and system for sending data across a UWBsignal using M-ary bi-orthogonal keying.

The following is a general description of a UWB system, notingparticularly how it is applicable to wireless networks. Although UWBtechnology has also been used in radar and ranging applications, thefollowing discussion addresses only issues relevant to wirelessnetworking applications.

It is helpful to briefly note some important design issues for indoorwireless networks. Such systems will need to operate over relativelyshort ranges in environments with multipath interference, but will needto provide high data rates, preferably using spectrum licensed by theFederal Communications Commission (FCC). Also, such systems are oftenused to support mobility, so they need low power dissipation to enablebattery operation and, as always, low cost and complexity is anadvantage.

Characteristics of UWB Systems

One embodiment of a UWB system uses signals that are based on trains ofshort duration pulses (also called chips) formed using a single basicpulse shape. The interval between individual pulses can be uniform orvariable, and there are a number of different methods that can be usedfor modulating the pulse train with data for communications. One commoncharacteristic in this embodiment, however, is that the pulse train istransmitted without translation to a higher carrier frequency, and soUWB transmissions using these sorts of pulses are sometimes also termed“carrier-less” radio transmissions. In other words, in this embodiment aUWB system drives its antenna directly with a baseband signal.

Another important point common to UWB systems is that the individualpulses are very short in duration, typically much shorter than theinterval corresponding to a single bit, which can offer advantages inresolving multipath components. We can represent a general UWB pulsetrain signal as a sum of pulses shifted in time, as shown in Equation 1:$\begin{matrix}{{s(t)} = {\sum\limits_{k = {- \infty}}^{\infty}{a_{k}{p( {t - t_{k}} )}}}} & (1)\end{matrix}$

Here s(t) is the UWB signal, p(t) is the basic pulse shape, and a_(k)and t_(k) are the amplitude and time offset for each individual pulse.Because of the short duration of the pulses, the spectrum of the UWBsignal can be several gigahertz or more in bandwidth. An example of atypical pulse stream is shown in FIG. 1. Here the pulse is a Gaussianmonopulse with a peak-to-peak time (T_(p-p)) of a fraction of ananosecond, a pulse period T_(p) of several nanoseconds, and a bandwidthof several gigahertz.

UWB Systems Limited to Low Power Spectral Density

UWB systems in general have extremely wide absolute bandwidth relativeto most existing wireless systems. This bandwidth is a directconsequence of the use of sub-nanosecond pulses that leads to signalbandwidths of several gigahertz or more. Because these signals are alsotransmitted without translation to higher center frequencies, it isclear that these signals will occupy the same frequency bands that arealready in use by many existing spectrum users.

Because of rulings by the FCC, future UWB systems will likely be limitedto operations using extremely low power spectral density (as measured indBm/MHz). Based on this fact, it is clear that even with a bandwidth ofseveral gigahertz, UWB systems will also be limited to relatively lowtotal transmit power. For example, a UWB system with 5 GHz of bandwidthmight have a maximum total transmit power of only a small fraction of amilliwatt over the entire 5 GHz of bandwidth.

Operation in the Power-Limited Regime

The bandwidth efficiency of a digital modulation scheme that transmits Bbits in T seconds (R bits/sec) using a bandwidth of W hertz is given byR/W=B/(WT) bits/s/Hz. As we will see, the bandwidth efficiency of a UWBsystem is not important in the sense of how efficiently it usesspectrum, but rather the value of this ratio serves to distinguish UWBsystems from more typical narrowband systems. Based on this ratio, R/W,digital communications systems can be classified as operating in eitherthe bandwidth-limited regime or the power-limited regime of thebandwidth-efficiency plane. This classification has fundamentalimplications for many of the important trade-offs that must be made inthe design of efficient communications systems.

For future UWB systems, the R/W ratio will likely be very low for thesystem to have any useful range. For example, even for a relativehigh-rate wireless network (say 100 Mbps), the bandwidth efficiency of aUWB wireless network will be as low as 1/20 or even 1/50, depending onthe bandwidth W. The primary consequence of this low value for the ratioR/W is that UWB systems will almost certainly operate well within thepower-limited regime of the bandwidth-efficiency plane.

The Critical Importance of Power Efficiency

The main result of UWB operation in the power-limited regime is thatsuch systems will be very sensitive to design issues that affect thepower efficiency of the system. For this reason, the analysis in thefollowing sections will focus on the critical issues of power efficiencyof the UWB modulation techniques, as well as the spectral effects ofmodulation that might also affect allowable transmit power levels. Theimplications of power-limited operation will also influence system-leveltrade-offs between range and data rate, as well as trade-offs betweencomplexity and performance in the form of forward error-correction.

Multipath Robustness and Precision Ranging

One frequently mentioned benefit of ultra-wide bandwidth is a robustnessto the effects of multipath interference. Multipath interference resultswhen multiple time-displaced copies of a signal reach a receiver at thesame time because of signal bounces in a cluttered environment. Thisrobustness is a result of two distinct factors: (1) wide fractionalbandwidth leads to less severe multipath fading, which is particularlyimportant for low-power wireless systems; and (2) wide absolutebandwidth enables resolution of multipath components and constructiveuse of multipath.

The effect of reduced multipath fading can be partially understood froma frequency-domain perspective by realizing that the absolute signalbandwidth of the UWB signal is much greater than the coherence bandwidthof nearly any conceivable multipath channel. Any frequency-selectivefades due to multipath will only affect a small portion of the signalpower for any channel realization. Previous work provides empiricalevidence that UWB signals experience a much lower variance in receivedsignal power in the presence of multipath than do narrowband signals.

For UWB signals, robustness to multipath fading is result not just ofthe wide system bandwidth, however, but is also a result of the largeratio of system bandwidth to center frequency, i.e., the fractionalbandwidth. A large fractional bandwidth means that there is acorresponding large variation in the mode and degree of RF energyinteraction with the surrounding environment over the entire UWBbandwidth. Environmental interactions such as scattering, refraction andreflection depend on the wavelength of the RF signals, and so the largefractional bandwidth leads to relatively low correlation in the fadingproperties of the different regions of the UWB bandwidth. Thus, theproperties of UWB signals should lead to more robust multipathperformance even than systems with equal bandwidth but much highercenter frequencies (i.e. lower fractional bandwidths).

The wide absolute bandwidth of UWB signals also provides fine timeresolution that enables a receiver to resolve and combine individualmultipath components, avoiding destructive interference.

Analysis of UWB Modulation Choices

Under current FCC regulations, UWB transmit power is limited by thepower spectral density (PSD) of the transmitted signal. FIG. 2 is agraph showing the power spectral density limits currently put in forceby the FCC.

This limitation affects the selection of a UWB modulation scheme in twodistinct ways. First, the modulation technique needs to be powerefficient. In other words, the modulation needs to provide the besterror performance for a given energy per bit. Second, the choice of amodulation scheme affects the structure of the PSD in the sense that itaffects the distribution of signal power over different frequency bands.If a particular modulation scheme results in the concentration of signalpower in narrow frequency ranges, it has the potential to imposeadditional constraints on the total transmit power in order to satisfythe PSD limitations.

As we compare different modulation schemes, we therefore examine boththe power efficiency and the effect of the modulation on the PSD. In thesections that follow, we examine a number of modulation schemes thathave been proposed for UWB, including several forms of pulse amplitudemodulation (PAM), such as: positive pulse amplitude modulation (PPAM),on-off keying (OOK), and binary phase-shift keying (BPSK), as well aspulse-position modulation (PPM).

Pulse Amplitude Modulation

As noted above, one general form of a UWB signal is a simple pulsetrain. Assuming that pulses are uniformly spaced in time (i.e. thek^(th) pulse occurs at time t=kT), then we can simplify Equation (1) to:$\begin{matrix}{{s(t)} = {\sum\limits_{k = {- \infty}}^{\infty}{a_{k}{p( {t - {kT}} )}}}} & (2)\end{matrix}$

where T is the pulse-spacing interval. From this general form of PAM, wecan analyze several specific modulation techniques by choosing themapping from data bits to pulse weights (a_(k)) in different ways. Thesedifferent techniques are illustrated in FIGS. 3A-3C, and are describedin the following paragraphs.

FIGS. 3A-3C are graphs showing exemplary pulse streams for OOK, PPAM,and BPSK modulation schemes, respectively. In each case, they show adata sequence “1 0 1 0.” FIGS. 4A-4C are constellation diagrams for themodulation schemes of FIGS. 3A-3C, respectively. As shown in FIGS.4A-4C, the constellation diagrams for OOK, PPAM, and BPSK are allone-dimensional, differing only in the symbol constellation's positionrelative to the origin.

On-Off Keying

As shown in FIG. 3A, OOK defines the data by the presence or absence ofa pulse. A “1” is indicated by a pulse, and a “0” is indicated by theabsence of a pulse. Thus, the bit stream “1 0 1 0” is indicated by thesequence of: a pulse, a blank where a pulse should be, a pulse, andanother blank.

This embodiment has a_(k)ε{0,2}, i.e., data bits are transmitted byeither the presence or absence of a pulse at time t=t_(k). In theconstellation diagram in FIG. 4A, this results in symbol points at (0,0)and (2,0).

Positive Pulse Amplitude Modulation

As shown in FIG. 3B, PPAM defines the data by the amplitude of thepulse. A “1” is indicated by a large pulse, and a “0” is indicated by asmall pulse. Thus, the bit stream “1 0 1 0” is indicated by the sequenceof: a large pulse, a small pulse, a large pulse, and a small pulse.

This embodiment uses strictly positive values for the two pulse weights,so that a_(k)ε{α₀,α₁} where 0<α₀<α₁. This corresponds to transmittingeither a large or small amplitude pulse based on the value of the sourcebit. In the constellation diagram of FIG. 4B this is shown as havingsignal points at (α₀, 0) and (α₁, 0).

Binary Phase Shift Keying

As shown in FIG. 3C, BPSK defines the data by the polarity of the pulse.A “1” is indicated by a non-inverted pulse, and a “0” is indicated by aninverted pulse. Thus, the bit stream “1 0 1 0” is indicated by thesequence of: a non-inverted pulse, an inverted pulse, a non-invertedpulse, and an inverted pulse.

In this embodiment a_(k)ε{−1,+1}. This corresponds to transmittingeither a non-inverted or an inverted pulse based on the value of thesource bit. In the constellation diagram of FIG. 4C this is shown ashaving signal points at (−1, 0) and (1, 0).

Pulse-Position Modulation

One other technique proposed for UWB pulse modulation, PPM, isfundamentally different from the PAM techniques described above becausethe pulses are not uniformly spaced in time. Rather, the source databits are used to modulate the time position of the individual pulsesinstead of the pulse amplitudes. For example, binary PPM encodes thedata bits in the pulse stream by advancing or delaying individual pulsesin time relative to uniform reference positions. In this case, theequation for the UWB signal becomes $\begin{matrix}{{s(t)} = {{\sum\limits_{k = {- \infty}}^{\infty}{p( {t - t_{k}} )}} = {\sum\limits_{k = {- \infty}}^{\infty}{p( {t - {kT} + {a_{k}\beta}} )}}}} & (3)\end{matrix}$

Here the data bits are mapped to the direction of the time shifts,a_(k), where a_(k)ε{−1,1}, and β is the amount of pulse advance or delayin time relative to the reference (unmodulated) position. When weconsider the constellation diagram for binary PPM, we find that the plotis no longer one dimensional, as it is for the binary PAM techniques.For PPM, the presence of two pulse with different time offsets resultsin a two-dimensional constellation plot. To find the specific locationof the symbol points within the plot, however, we need to determine thecorrelation ρ between the two different symbols, the advanced anddelayed pulses. $\begin{matrix}{\rho = \frac{\int_{= {- \infty}}^{\infty}{{p( {t - {\beta\quad T}} )}{p( {t + {\beta\quad T}} )}\quad{\mathbb{d}t}}}{\int_{= {- \infty}}^{\infty}{{p(t)}{p(t)}{\mathbb{d}t}}}} & (4)\end{matrix}$

FIGS. 5A-5C are constellation diagrams for pulse position modulationschemes under various conditions for binary PPM based on the pulse shownin FIG. 1. FIG. 5A shows as situation where the pulses are orthogonal(i.e., ρ=0); FIG. 5B shows the situation where the pulses are notorthogonal and ρ>0; and FIG. 5C shows the situation where the pulses arenot orthogonal and ρ<0.

For the non-orthogonal cases of binary PPM, the orthogonal basisfunction used to define the constellation plot can be found usingGram-Schmidt orthogonalization for the two non-orthogonal pulses. Theconstellation diagrams in FIGS. 5A-5C all have symbol points at (1,0)and (ρ,√{square root over (1−ρ²)}), and the two symbol points lie on theunit circle (when normalized to unit energy).

In the case where the two different locations of the pulse have nooverlap in time, the correlation will clearly be (ρ≈0) and the binaryPPM becomes orthogonal modulation. The constellation for this case isshown in FIG. 5A, where the symbol points are (1,0) and (0,1). Here thetwo orthogonal pulses have been used to create orthogonal basis vectorsfor the constellation plot.

When the two pulses overlap, the correlation ρ in general will not bezero, but will range between one and some minimum (possibly negative)value.

Comparison of Power Efficiency for Binary Modulation

We can use the constellation diagrams in FIGS. 4A-4C and 5A-5C tocompare the power efficiency of the various binary modulation techniquesby computing the inter-symbol distance, d, as a function of averagesymbol energy, E_(s). For OOK, we have${E_{s} = \frac{( {0^{2} + d^{2}} )}{2}},{{{so}{\quad\quad}d} = {\sqrt{2E_{s}}.}}$For positive-valued PAM (PPAM) we see that${d = ( {\alpha_{1} - \alpha_{0}} )},{{{so}\quad E_{s}} = {\frac{( {\alpha_{0}^{2} + ( {d + \alpha_{0}} )^{2}} }{2}.}}$Solving for d, we get d=(√{square root over (2E_(s)−α₀ ²)}−α₀). If weassume α₀≧0, then we have d≦√{square root over (2E_(S))}, which issatisfied with equality when α₀=0 (i.e. when PPAM becomes OOK). Forantipodal binary PAM (BPSK) we have${E_{s} = ( \frac{d}{2} )^{2}},{{{so}\quad d} = {2{\sqrt{E_{s}}.}}}$

For binary PPM, the inter-symbol distance depends on the correlationbetween the advanced and delayed pulses defined in Equation (4) and forthe general case, d=√{square root over (2E_(S)(1−ρ))}. Here we see thatif the value of ρ ranges between −1 and +1, the distance can rangebetween d=0 and d=2√{square root over (E_(s))}. The actual maximum andminimum values for ρ that determine this range of possible inter-symboldistances depend on the specific shape of the pulse p(t) and can bedetermined according to Equation (4) for different values of β. For theexample Gaussian monopulse shown in FIG. 1, the value of ρ as defined inEquation (4) ranges from (+1) to approximately (−0.45) as β ranges fromzero to several multiples of T_(p). TABLE 1 Differences BetweenModulation Techniques Power Efficiency Modulation Inter-symbol Relativeto Class Specific Form Distance Antipodal Signaling Pulse- Orthogonal$d = \sqrt{2E_{b}}$ −3 dB position Non-orthogonal$d = \sqrt{2{E_{b}( {1 - \rho} )}}$ <1.4 dB (variable)Modulation Amplitude Positive PAM $d < \sqrt{2E_{b}}$ <−3 dB  ModulationOOK $d = \sqrt{2E_{b}}$ −3 dB Antipodal $d = {2\sqrt{E_{b}}}$  0 dB

These results show significant differences between the modulationtechniques and are summarized in Table 1. The orthogonal PPM and OOKtechniques are equally efficient and the positive PAM system is less so,but becomes the same in the limit as the PAM becomes OOK. Non-orthogonalPPM has a power efficiency that depends on the symbol correlation ρ, butis still suboptimal. Antipodal signaling (BPSK) provides the greatestinter-symbol distance for a given average symbol energy. This differenceprovides at least a 3 dB advantage in efficiency relative to OOK, PPAM,or orthogonal PPM, and to achieve the same bit error rate (which is afunction of distance) PPM or OOK must use double the bit energy, or 3 dBhigher E_(b).

Decomposition of Binary Modulation Techniques

For the binary PAM techniques depicted in FIGS. 4A-4C, the constellationdiagrams differ only in their position relative to the origin. It is awell-known result in communications theory that power efficiency dependson the mean of the symbol constellation—this is why the zero-meanproperty of the BPSK makes it superior in the ratio of inter-symboldistance to symbol energy. Another way to understand this difference isto decompose the weight sequence into a constant value sequence added toa zero-mean random sequence: a_(k)=μ_(a)+z_(k). This sequencedecomposition allows us to represent the UWB pulse train as the sum ofan unmodulated component pulse train and an antipodal component pulsetrain: $\begin{matrix}{{s(t)} = {{\sum\limits_{k = {- \infty}}^{\infty}{\mu_{a}{p( {t - {kT}} )}}} = {\sum\limits_{k = {- \infty}}^{\infty}{z_{k}{p( {t - {kT}} )}}}}} & (5)\end{matrix}$

From this result we can easily see the source of the difference in powerefficiency for the PAM techniques. The energy in the unmodulatedcomponent of the pulse train above does not contribute to communicatingdata between the transmitter and receiver, and is effectively wasted.Only the energy in the antipodal component contributes to thecommunications process. The greater the energy in the unmodulatedcomponent (i.e. the higher the mean μ_(a) for a give distance d) thepoorer is the power efficiency of the modulation. BPSK is thus seen tobe optimal for binary techniques since it has zero-mean and all of itsenergy is contained in the antipodal component of the pulse train.

For PPM, we can perform a similar, but more general, decomposition ofthe pulse train. Here we must use unmodulated and antipodal componentsthat, unlike PAM, have different pulse shapes. We define two new pulses:$\begin{matrix}{{{m(t)} = \frac{{p( {t - {\beta\quad T}} )} + {p( {t + {\beta\quad T}} )}}{2}},{{{and}\quad{b(t)}} = \frac{{p( {t - {\beta\quad T}} )} - {p( {t + {\beta\quad T}} )}}{2}}} & (6)\end{matrix}$

These pulses represent the unmodulated [m(t)] and antipodal [b(t)] pulsetrain components. We can use these two pulses to write the UWB pulsetrain as the sum of two separate component pulse trains: $\begin{matrix}{{s(t)} = {{\sum\limits_{k = {- \infty}}^{\infty}{\mu_{a}{m( {t - {kT}} )}}} + {\sum\limits_{k = {- \infty}}^{\infty}{z_{k}{b( {t - {kT}} )}}}}} & (7)\end{matrix}$

Using this decomposition, we see that data bits are transmitted bysending either [m(t)+b(t)] or [m(t)−b(t)] at each time interval t=kT.The sign of the component m(t) is independent of the data value and istherefore not modulated. Two examples of this decomposition for binaryPPM are shown in FIGS. 6A and 6B for values of β that result in bothoverlapping and non-overlapping pulses.

FIGS. 6A-6D are graphs showing component pulses for the decomposition ofbinary PPM into unmodulated and antipodal pulse trains. FIG. 6A showsthe original pulses with β=5T_(p); FIG. 6B shows the original pulseswith β=1.5T_(p); FIG. 6C shows the unmodulated component pulse [m(t)]and antipodal component pulse [b(t)] for β=5T_(p); and FIG. 6D shows theunmodulated component pulse [m(t)] and antipodal component pulse [b(t)]for β=1.5T_(p).

As with the PAM cases above, we can see that the energy in theunmodulated component of the pulse train defined by m(t) is useless inthe communication of information and leads to inefficient modulation.

Spectral Effects of Modulation Techniques

Another important consideration in evaluating a UWB modulation techniqueis the effect of the modulation on the spectrum of the transmittedsignal. As noted in an earlier section, UWB signals have been limited bythe FCC by the peak of their PSD, so that for best system performancesignals should be designed to maximize transmit power for given limitson PSD levels.

Spectral Analysis for PAM

To understand the effect of the modulation scheme on the UWB signal, weneed to find the spectrum not of the isolated pulse, but of themodulated pulse train. If we assume that the modulating data are random,the transmitted pulse train is also a random signal and as such does nothave a deterministic Fourier transform. However, we can still understandthe effects of modulation on the spectral distribution of signal powerby finding its expectation over the random source data sequences. Thispower spectral density (PSD) of the transmit signal, s(t), is theFourier transform of the signal autocorrelation and is denoted byΦ_(SS)(f). Because the pulses in a PAM UWB signal are uniformly spacedas in Equation (2), we can derive a general form for the PSD of the PAMsignals as follows:Φ_(SS)(f)=|P(f)|²Φ_(aa)(f)  (8)

Here P(f) is the Fourier transform of the basic pulse, p(t), andΦ_(aa)(f) is the PSD of the random data sequence, a_(k), which ishereafter assumed to be a wide-sense stationary random sequence. If weassume that the pulse weights a_(k) correspond to the data bits to betransmitted and that the random data are independent and identicallydistributed (IID), then the PSD can be determined as follows:$\begin{matrix}{{\Phi_{aa}(f)} = {\sigma_{a}^{2} + {\frac{\mu_{a}^{2}}{T}{\sum\limits_{k = {- \infty}}^{\infty}{\delta( {f - \frac{k}{T}} )}}}}} & (9)\end{matrix}$

where σ_(a) ² and μ_(a) are the variance and mean of the weight sequenceand δ(f) is a unit impulse function. This PSD is periodic in thefrequency domain with period $f = \frac{1}{T}$because it is the transform of the discrete auto-correlation sequence,Φ_(aa)(k)=E{a_(n)+_(k)a_(n)*}. This PSD in Equation (9) has both acontinuous portion and discrete spectral lines, corresponding to thefirst and second terms on the right-hand side. It is worth noting thatthe magnitude of the spectral lines depends on the mean of the weights,μ_(a). In light of the decomposition described above, we see that theenergy in the unmodulated component of the pulse train is the energy inthe spectral lines and the energy in the antipodal component is theenergy of the continuous spectral component.

When we combine the results of Equations (8) and (9) we see that theresulting PSD of the transmitted signal is equivalent to the result offiltering a weighted impulse sequence through a filter with frequencyresponse P(f): $\begin{matrix}{{\Phi_{ss}(f)} = {{\frac{\sigma_{a}^{2}}{T}{{P(f)}}^{2}} + {\frac{\mu_{a}^{2}}{T^{2}}{\sum\limits_{k = {- \infty}}^{\infty}{{{P( \frac{k}{T} )}}^{2}{\delta( {f - \frac{k}{T}} )}}}}}} & (10)\end{matrix}$

At this point we can again consider the different modulation techniquesfor PAM described earlier. The PSD for the OOK signal with pulseamplitudes weights a_(k)ε{0,2} is determined as follows: $\begin{matrix}{{\Phi_{{ss},{OOK}}(f)} = {{\frac{1}{T}{{P(f)}}^{2}} + {\frac{1}{T^{2}}{\sum\limits_{k = {- \infty}}^{\infty}{{{P( \frac{k}{T} )}}^{2}{\delta( {f - \frac{k}{T}} )}}}}}} & (11)\end{matrix}$

In this equation we see that OOK results in discrete spectral lines inthe PSD of the UWB signal. The spectral lines are spaced at a frequencyinterval of $f = \frac{1}{T}$and each line has power proportional to P(f) evaluated at$F = {\frac{k}{T}.}$For OOK the total power in the spectral lines is equal to the power inthe continuous component of the PSD, as shown above. A similar result isobtained for the positive-valued PAM signal, where we have$\sigma_{a}^{2} = {{\frac{( {\alpha_{0} - \alpha_{1}} )^{2}}{4}\quad{and}\quad\mu_{a}} = {\frac{( {\alpha_{0} + \alpha_{1}} )}{2}.}}$Substituting these values in Equation (8) the PSD becomes:$\begin{matrix}{{\Phi_{{ss},{PPAM}}(f)} = {{\frac{( {\alpha_{0} - \alpha_{1}} )^{2}}{4T}{{P(f)}}^{2}} + {\frac{( {\alpha_{0} - \alpha_{1}} )^{2}}{4T^{2}}{\sum\limits_{k = {- \infty}}^{\infty}{{{P( \frac{k}{T} )}}^{2}{\delta( {f - \frac{k}{T}} )}}}}}} & (12)\end{matrix}$

We see that, as with OOK, there are spectral lines present in thetransmitted signal for positive-valued PAM and furthermore that themagnitude of the lines increases with the weight sequence mean. Notethat PPAM spectrum becomes the same as the OOK spectrum when α₀→0.

The situation is very different for antipodal signaling, wherea_(k)ε{−1,+1}, so that σ_(a) ²=1 and μ_(a)=0. In this case, the PSDbecomes simply: $\begin{matrix}{{\Phi_{{ss},{BPSK}}(f)} = {{\frac{\sigma_{a}^{2}}{T}{{P(f)}}^{2}} = {\frac{1}{T}{{P(f)}}^{2}}}} & (13)\end{matrix}$

Here we see that the spectral lines vanish because of the zero mean ofthe weight sequence. Because the PSD for BPSK has no lines, the spectraldistribution of energy does not depend on the pulse interval T or thepulse-repetition frequency (PRF). Rather the presence of T in Equation(13) only shows that the total power of the transmit signal increaseslinearly at all frequencies with the PRF when pulse amplitude isconstant.

Spectral Analysis for PPM

The results of Equations (8) and (9) do not directly apply to the caseof PPM because the pulses do not have uniform spacing in time. To findthe PSD for PPM signals, however, we can use the decomposition techniquedescribed in Equation (7) above that allowed us to represent the PPMsignal as the sum of two uniformly spaced pulse trains. From thedefinitions in Equation (6) it is clear that m(t) and b(t) areorthogonal regardless of the orthogonality of the shifted pulses p(t−β)and p(tβ). Using this fact, we can find the PSD of the composite pulsetrain in Equation (7), the PSD of the binary PPM signal as follows:$\begin{matrix}{{\Phi_{{ss},{PPM}}(f)} = {{\frac{\sigma_{a}^{2}}{T}{{B(f)}}^{2}} + {\frac{\mu_{a}^{2}}{T^{2}}{\sum\limits_{k = {- \infty}}^{\infty}{{{M( \frac{k}{T} )}}^{2}{\delta( {f - \frac{k}{T}} )}}}}}} & (14)\end{matrix}$

Where B(f) and M(f) are the Fourier transforms of the component pulsesb(t) and m(t), respectively. As with the case of the PAM signals, it isclear that the energy that corresponded to the unmodulated pulse trainin Equation (7) here translates to energy contained in spectral lines.Similarly, the energy in the antipodal portion of the signal translatesto the energy of the continuous spectral component of Equation (14).

One significant difference between the PAM and PPM spectra is that forPPM the envelope of the magnitudes of the spectral lines can bedifferent from the shape of the continuous spectrum.

The continuous component of the PSD has a shape that depends on B(f),but the power distribution in the spectral lines depends on M(f). Thesespectral lines still have a frequency spacing of ${f = \frac{1}{T}},$but the distribution of power in the lines can be significantlydifferent.

In general, both the distribution of energy between the discrete andcontinuous components of the spectrum, as well as the distribution ofspectral energy with respect to frequency, depend on the shape of theoriginal pulse p(t) and the magnitude of the time shift, βT. As with thePAM signals, we can conclude that from the viewpoint of the systemdesigner it is desirable to minimize the energy in the spectral lines.For PPM this is done by minimizing the correlation value ρ, with theadditional consideration that the shape of the component pulses m(t) andb(t) may result in less uniform distribution of energy in the spectrum.This could in turn lead to suboptimal designs for a PSD-limited system.

SUMMARY OF THE INVENTION

Consistent with the title of this section, only a brief description ofselected features of the present invention is now presented. A morecomplete description of the present invention is the subject of thisentire document.

An object of the present invention is to provide multiple acquisitionmodes to maximize the transmission time of a network based on thecurrent signal quality.

Another object of the present invention is to provide a way in which adevice can switch between these multiple modes in an efficient manner.

These and other objects are accomplished by way of a method of acquiringincoming signals in a wireless network device, comprising: acquiring theincoming signals using a default acquisition preamble having a defaultsynchronization period; evaluating the incoming signals to determinewhether first signal parameters are met; changing to a first alternateacquisition preamble having a first alternate synchronization period ifthe first signal parameters are met; evaluating the incoming signals todetermine whether second signal parameters are met; and changing to asecond alternate acquisition preamble having a second alternatesynchronization period if the second parameters are met, wherein thedefault synchronization period, the first alternate synchronizationperiod, and the second alternate synchronization period all havedifferent values.

The default acquisition preamble may be a normal preamble having anormal synchronization period, with the first acquisition alternatepreamble being a short preamble having a short synchronization period,and the second alternate acquisition preamble being a long preamblehaving a long synchronization period. The normal synchronization periodis preferably longer than the short synchronization period, and the longsynchronization period is preferably longer than the normalsynchronization period.

The first signal parameters may be met if the signal strength is below afirst threshold, and the second signal parameters may be met if thesignal strength is above a second threshold. In the alternative, thefirst signal parameters may be met if a number of requested packetretransmissions per unit time in the wireless network device is below afirst threshold, and the second signal parameters may be met if thenumber of requested packet retransmissions per unit time in the wirelessnetwork device is above a second threshold.

The normal acquisition preamble preferably includes a normal decisionfeedback equalization training period, the long acquisition preambleincludes a long decision feedback equalization training period, and thelong decision feedback equalization training period is preferably longerthan the normal decision feedback equalization training period.

The default acquisition preamble may also be a short preamble having ashort synchronization period, with the first acquisition alternatepreamble being a normal preamble having a normal synchronization period,and the second alternate acquisition preamble being a long preamblehaving a long synchronization period. The normal synchronization periodis preferably longer than the short synchronization period, and the longsynchronization period is preferably longer than the normalsynchronization period.

The first signal parameters may be met if the signal strength is below afirst threshold but not below a second threshold, and the second signalparameters may be met if the signal strength is below both the first andsecond thresholds. In the alternative, the first signal parameters maybe met if a number of requested packet retransmissions per unit time inthe wireless network device is below a first threshold but not below asecond threshold, and the second signal parameters may be met if thenumber of requested packet retransmissions per unit time in the wirelessnetwork device is below both the first and second thresholds.

The normal acquisition preamble preferably includes a normal decisionfeedback equalization training period, the long acquisition preamblepreferably includes a long decision feedback equalization trainingperiod, and the long decision feedback equalization training period ispreferably longer than the normal decision feedback equalizationtraining period.

The default acquisition preamble may be a long preamble having a longsynchronization period, with the first acquisition alternate preamblebeing a normal preamble having a normal synchronization period, and thesecond alternate acquisition preamble being a short preamble having ashort synchronization period. The normal synchronization period ispreferably longer than the short synchronization period, and the longsynchronization period is preferably longer than the normalsynchronization period.

The first signal parameters may be met if the signal strength is above afirst threshold but not above a second threshold, and the second signalparameters may be met if the signal strength is above both the first andsecond thresholds. In the alternative, the first signal parameters maybe met if a number of requested packet retransmissions per unit time inthe wireless network device is above a first threshold but not above asecond threshold, and the second signal parameters may be met if thenumber of requested packet retransmissions per unit time in the wirelessnetwork device is above both the first and second thresholds.

The normal acquisition preamble preferably includes a normal decisionfeedback equalization training period, the long acquisition preamblepreferably includes a long decision feedback equalization trainingperiod, and the long decision feedback equalization training period ispreferably longer than the normal decision feedback equalizationtraining period.

The wireless network device is preferably an ultrawide bandwidth device.

A method is also provided of acquiring incoming signals in a wirelessnetwork device, comprising: acquiring the incoming signals using adefault acquisition preamble having a default synchronization period;evaluating the incoming signals to determine whether i^(th) signalparameters are met; and changing to an i^(th) alternate acquisitionpreamble having an i^(th) synchronization period if the i^(th) signalparameters are met, wherein k is an integer greater than 1, wherein i isan integer that varies from 1 to k, wherein the first through k^(th)synchronization periods all have different values, and wherein the firstthrough k^(th) signal parameters are mutually exclusive.

The wireless network device is preferably an ultrawide bandwidth device.

BRIEF DESCRIPTION OF THE DRAWINGS

A more complete appreciation of the invention and its many attendantadvantages will be readily obtained as it becomes better understood withreference to the following detailed description when considered inconnection with the accompanying drawings, in which:

FIG. 1 is a graph of a typical UWB pulse stream;

FIG. 2 is a graph showing the power spectral density limits currentlyput in force by the FCC;

FIGS. 3A and 3B are graphs showing exemplary pulse streams for on-offkeying, positive pulse amplitude modulation, and binary phase-shiftkeying, respectively using monopulses, according to a preferredembodiment of the present invention;

FIGS. 4A-4C are constellation diagrams for the modulation schemes ofFIGS. 3A-3C, respectively;

FIGS. 5A-5C are constellation diagrams for pulse position modulationschemes under various conditions for binary pulse position modulationschemes, based on the pulse shown in FIG. 1;

FIGS. 6A-6D are graphs showing component pulses for the decomposition ofbinary PPM into unmodulated and antipodal pulse trains;

FIG. 7 is a timing diagram showing a one-pulse code word usingmonopulses according to a preferred embodiment of the present invention;

FIG. 8 is a timing diagram showing a five-pulse code word usingmonopulses, according to a preferred embodiment of the presentinvention;

FIG. 9 is a graph of a typical oscillating signal used to form a pulsestream in a preferred embodiment of the present invention;

FIGS. 10A-10C are graphs showing exemplary pulse streams for OOK, PPAM,and BPSK modulation schemes, respectively, using portions of anoscillating signal as pulses, according to preferred embodiments of thepresent invention;

FIG. 11 is a timing diagram showing a five-pulse code word using arepeated oscillating signal pulse, according to a preferred embodimentof the present invention;

FIGS. 12A and 12B are timing diagrams showing a five-pulse code wordusing five ternary pulses, according to preferred embodiments of thepresent invention;

FIG. 13 is a graph of a UWB PSD using a notch according to a preferredembodiment of the present invention;

FIG. 14 is a graph of a UWB transmission scheme using two bandsaccording to a preferred embodiment of the present invention;

FIGS. 15A and 15B are block diagrams of a transmitter and receiver pairaccording to preferred embodiments of the present invention;

FIG. 16A is a block diagram of the correlator of FIGS. 15A and 15Bhaving one arm according to a preferred embodiment of the presentinvention;

FIG. 16B is a block diagram of the correlator of FIGS. 15A and 15Bhaving two arms according to a preferred embodiment of the presentinvention;

FIG. 16C is a block diagram of the correlator of FIGS. 15A and 15Bhaving more than two arms according to a preferred embodiment of thepresent invention;

FIGS. 17A and 17B are block diagram showing a UWB system usingpseudo-random scrambling, according to preferred embodiments of thepresent invention;

FIG. 18 is a block diagram of a data packet according to a preferredembodiment of the present invention;

FIGS. 19 and 20 are flow charts describing the operation of thetransmitter and receiver, respectively, according to a preferredembodiment of the present invention;

FIGS. 21A and 21B are block diagrams showing circuits for performing arapid clear channel assessment according to preferred embodiments of thepresent invention;

FIGS. 22A-22C are block diagrams of a short preamble, a normal preamble,and a long preamble, respectively, according to preferred embodiments ofthe present invention; and

FIGS. 23A and 23B are graphs showing the output of the squarer elementsin FIGS. 21A and 21B for three and seven term squaring, respectively,according to preferred embodiments of the present invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Preferred embodiments of the present invention will now be describedwith reference to the drawings. Throughout the several views, likereference numerals designate identical or corresponding parts.

Binary Systems

As noted above with respect to FIG. 1, in one embodiment of a UWBsystem, a series of pulses are sent across a transmission medium. Inorder to carry data, these UWB pulses need to have data encoded (i.e.,modulated) into them. Then a receiver can look at the incoming pulsesand decode the original data. As noted above with respect to FIGS. 3A-3Cand 6A-6D, a number of different approaches have been tried, includingvarious PAM and PPM schemes.

PPM shifts the position of individual pulses depending upon whether thepulse needs to represent a “1” or a “0.” As shown, for example, in FIG.6A, in a simple PPM scheme a pulse is moved from a default position by adistance βT to the left if it represents a “0” and is moved from thedefault position by a distance βT to the right if it represents a “1.”

In this method, the pulses don't change, they just advance or delay intime, i.e., the position of these pulses is modulated in time. In fact,the pulses are generally identical, which makes it easier to generatethem. In FIG. 6A the pulses all rise first and then fall.

As noted with respect to FIG. 3C, BPSK does not shift the position ofthe pulses, but rather inverts the pulses to pass data. As shown in theembodiment of FIG. 3C, a pulse is unaltered if it represents a “0” andis inverted if it represents a “1.” In either case the position of thepulse remains unchanged.

In most cases, BPSK signals will be superior to PPM signals. One primaryreason is how the two methods handle noise. When a signal gets sent froma transmitter to a receiver, it is subjected to a certain amount ofnoise. This noise rides on top of the data signal and can distort thesignal. Some of the noise comes from going through the channel (i.e.,the transmission medium). Additional noise comes from the receiver,which has to amplify a very small signal. Such an amplification processinherently introduces noise.

The way to compare individual transmission schemes is to determine themaximum amount of noise allowable before the system exceeds a maximumerror rate, In any transmission system some errors will occur, due tonoise and other reasons. A given system will set a maximum allowableerror rate, which it is designed to compensate for. Beyond this errorrate, the system will not achieve a desired level of performance. Anexemplary maximum error rate, often called a bit error rate (BER), isone error in a thousand, often described as having a BER of 10^(−3.)

In the same noise environment, a PPM signal will require twice as muchtransmit power to achieve the same BER as a BPSK signal. Another way ofsaying this is that the BPSK signal is superior to the PPM signal by 3dB (i.e., by a factor of two in power). In other words, for the sameamount of power, the BPSK signal will tolerate more noise than a PPMsignal. And to tolerate the same amount of noise, the PPM signal wouldrequire more power than the BPSK signal.

This analysis assumes Gaussian noise. If the noise were non-Gaussian,the benefits of a BPSK signal might vary (either higher or lower), ormight remain the same.

An alternate transmission scheme would be pulse amplitude modulation(PAM), which encodes data through the use of pulses with differentamplitude, as described with respect to FIG. 3B. As shown in theembodiment of FIG. 3B, a small pulse is sent to represent a “0” and alarge pulse is sent to represent a “1.” Regardless, the pulsespreferably remain a standard distance from each other.

These are all binary systems, i.e., they encode data as a string of onesand zeroes.

M-ary Systems

The analysis in the Background of the Invention was restricted to binarymodulation techniques appropriate for UWB signal generation. There arealso a number of ways in which these binary techniques could be extendedto larger symbol constellations. A few specific exemplary forms arediscussed below.

Such alternative transmission schemes may be called M-ary systems. Inthis case, M-ary simply means that there are M different choices forencoding data. A binary system is actually an M-ary system where (M=2),i.e., a 2-ary system. Alternate systems could have M equal to four(4-ary), M equal to eight (8-ary), or any other acceptable number.Powers of two are preferable for M since it makes implementation easier,but are not required.

In an M-ary system, each pulse has M different ways that it can be sentto the receiver. For example an M-ary PAM system (called an MPAM system)would have M different pulse voltages that can be used.

An analysis of M-PAM shows that higher order PAM modulation leads toless power efficiency. This is clearly seen by the reduced data ratesversus range performance for the larger constellations. However, such adesign might be more robust against multipath-induced inter-symbolinterference (ISI) due to the longer symbol interval for a given datarate.

The binary PPM technique can also be extended to M-ary orthogonal (ornon-orthogonal) PPM by mapping b bits to a single pulse (or pulse train)and using 2^(b) different values for the pulse position. In generalM-ary orthogonal signaling will provide better distance properties forhigher dimensions, resulting in better power efficiency relative tobinary PPM. M-ary PPM can be analyzed by extending the decompositiontechniques describe earlier for the binary case. It is known that M-aryorthogonal constellation do have non-zero means and this technique wouldtherefore still result in spectral lines and suboptimal powerefficiency.

The most promising extension of binary modulation is to map multipledata bits into bi-orthogonal sequences of bi-phase pulses. This system,called M-ary bi-orthogonal keying (MBOK) involves the mapping of b bitsto a group of consecutive bi-phase pulses. MBOK provides improved powerefficiency relative to binary antipodal signaling, yet would still notgenerate spectral lines for white data.

In addition, some of the transmission schemes set forth above could becombined into a single M-ary system. An M-ary BPSK PPAM system could useM/2 voltages, with the pulses being either non-inverted or inverted toachieve M possible results. Such a system can be referred to as simplyM-PAM, e.g., 8-PAM or 4-PAM. The term M-PAM is generally limited tocases where M>2, since a 2-PAM system would be equivalent to a basicBPSK system.

UWB systems preferably transmit at extremely small power levels, but atvery wide bandwidths. Thus, although they generally have essentially asmuch bandwidth as they want, UWB systems must maintain very low powerlevels to be efficient. Therefore, it's desirable to choose a modulationscheme that is extremely power efficient.

It turns out that M-PAM modulation is less power efficient than BPSKmodulation, for example, consider an 8-PAM transmission scheme. Althoughit appears that such a scheme would be more efficient (after all, it'stransmitting three times as much data in a similar transmission usingBPSK), it turns out to be less power efficient than BPSK. The reason forthis is that an M-PAM modulation scheme requires much higher powerlevels for many of its pulses.

However, there are more power efficient alternatives for using a BPSKsignal at a given frequency, (i.e., number of pulses per unit time) tosend more data.

Increased Frequency

In one embodiment it is possible to simply increase the frequency of thetransmitted pulses, (i.e., send more pulses per unit time), rather thanuse an M-PAM (or other) modulation scheme. This will be effective up tothe maximum possible transmission frequency of the UWB signal, i.e., thefastest the system can send pulses.

Therefore, the important questions are: how fast can you run the systemclock; and how fast can you sent pulses? In many UWB systems the pulsesare on the order of one nanosecond in width. In order to send onehundred megabits per second (MBPS) of data, the system must send one bitof data every ten nanoseconds

Thus, for a transmission rate of 100 MBPS with each bit of datarepresented by a single pulse, the system need only send a single 1 nspulse every 10 ns—a reasonable requirement. This can potentially allowfor more pulses to be sent, thus increasing the rate of datatransmission. However, this is limited by the size of the pulse and theminimum allowable distance between pulses. Once the pulses are so closethat they are only that minimum distance from each other, the system canno longer increase the pulse transmission rate without having pulsescollide.

Code Word

An alternative to sending data as individual pulses is to insteadrepresent each bit by a series of pulses. This series of pulses can becalled a code word. In a binary system, a set of BPSK pulses willpreferably be chosen to represent a “0” and its inverse will preferablybe chosen to represent a “1.”

Individual pulses are then ordered together into code words to transferdata at a given data rate, with each code word corresponding to one ormore bits of information to be transferred. The code words have a codeword period T_(cw), indicating the duration of an code word, and arelated code word frequency F_(cw). This may correspond to the datarate, though it does not have to. FIGS. 7 and 8 show two examples ofcode words.

FIG. 7 is a timing diagram showing a one-pulse code word usingmonopulses according to a preferred embodiment of the present invention.This simplest example has a code word that includes a single pulse. Inthis case the code word period T_(cw) and the pulse period T_(p) are thesame (i.e., the pulses and the code words are transmitted at the samefrequency). As shown in FIG. 7, the non-inverted pulse corresponds to a“1,” and the inverted pulse corresponds to a “0.” This could be reversedfor alternate embodiments.

FIG. 8 is a timing diagram showing a five-pulse code word usingmonopulses according to a preferred embodiment of the present invention.This embodiment has a code word that includes five binary pulses. Inthis case the code word period T_(cw) is five times the pulse periodT_(p) (i.e., the code words are transmitted at one-fifth the frequencyof the pulses).

In other words:T _(cw) =n*Tp  (15)

for an n-pulse code word. Thus, the pulse period T_(p) and number ofpulses n per code word determine the period of the code word T_(cw).

As shown in FIG. 8, a particular orientation of the five pulsescorresponds to a “1,” and the inverse of this orientation corresponds toa “0.” The particular choice of pulse orientation and arrangement withinthe code word is not critical, and can be varied as necessary. What isimportant is that the “1” and “0” code words are the inverse of eachother.

One preferred embodiment includes 13 analog pulses per code word, andsets the pulse frequency F_(p) at 1.3 GHz (770 ps pulse period T_(p)).This results in a code word frequency F_(cw) of 100 MHz (10 ns code wordperiod T_(cw)), which corresponds to a data transfer rate of 100 Mbitsof information per second. Another preferred embodiment includes 24analog pulses per code word, and sets the pulse frequency F_(p) ateither 1.368 GHz or 2.736 GHz (731 ps or 365.5 ps pulse period T_(p))for each of two bands used. (This is a multiple band approach, as willbe described below.) This results in a code word frequency F_(cw) of 57MHz or 114 MHz (17.54 ns or 8.77 ns code word period T_(cw)), dependingupon the band.

The various parameters of peak-to-peak pulse width T_(p-p), pulse periodT_(p), pulse frequency F_(p), number of pulses per code word n, codeword period T_(cw), and code word frequency F_(cw) can be varied asnecessary to achieve the desired performance characteristics for thetransceiver. For example, the embodiments disclosed in FIGS. 7 and 8have the same code word period T_(cw), despite the differing number ofpulses n. This means that the transmission power for a given code wordperiod T_(cw) is used in a single pulse in the embodiment of FIG. 7, butis spread out over five pulses in the embodiment of FIG. 8. Alternateembodiments can obviously change these parameters as needed.

Thus, in the embodiment of FIG. 7, when a transmitter passes a bit ofdata to a receiver, the transmitter sends the bit as a code word (i.e.,a set series of pulses). As noted above, the bits are preferablyrepresented by inverse code words such that the non-inverted code wordrepresents a “1” and the inverted code word represents a “1.” However,in alternate embodiments this assignment of code word/inverse code wordto “1” and “0” values can be reversed.

In addition, although FIG. 8 shows a code word having five pulses forthe sake of simplicity, this number can be varied as needed. In fact, asnoted above, 13 or 24 are preferred numbers of pulses per code word.Alternate embodiments can use any code word length that allows systemrequirements to be met. For example, as clock speeds increase, thenumber of pulses that can be sent in a given time will increase andlonger code word lengths may be used.

One advantage with using a code word is that you can spread out arequired transmission power over multiple pulses. For a successfultransmission, it's necessary to use a certain amount of energy to sendeach bit. If the bit is sent in a single pulse, that pulse has toinclude all of the required energy. This requires a larger pulse andincreases the peak-to-average ratio of the signal (i.e., the entirewaveform). However, if five pulses are used to send a single bit of data(as shown in the embodiment of FIG. 8), the energy can be spread outamong five separate pulses. Thus, each individual pulse can be smallerand can have a lower peak-to-average ratio.

However, using a multiple pulse code word to represent a bit will alterthe spectrum of the transmitted signal. And given rules currentlypromulgated by the Federal Communications Commission (FCC), UWB signalsmust meet certain power constraints. (See FIG. 2)

As shown in FIG. 2, UWB signals must fall below a set power maximum forany given frequency. In other words, the energy of the UWB signal cannotexceed the set power maximum at any frequency. Therefore, it isnecessary that the UWB signal fit within the power spectrum densityrequirements set forth in FIG. 2.

For a UWB signal, the amount of energy sent in a transmission is equalto the area under its power spectrum density curve. For the bestpossible system performance, it's preferable that this area bemaximized. In other words, it is desirable to have a signal whoseproperties fits under the restricted curve, but is arranged to have amaximum possible area.

When a UWB system uses a sequence of pulses to send a bit of data, thepower spectrum density ends up looking wavy, with numerous peaks andvalleys. The exact waviness of the power spectrum density depends uponthe particular sequence of pulses used.

The peaks in the PSD curve can limit the transmission power by limitingthe maximum total power transmitted. Since the power spectrum densitycannot ever go above the power maximum set by the FCC, the maximum pointof the power spectrum density curve can be no higher than the allowablepower maximum. If there are too many peaks (and corresponding valleys)in the power spectrum density curve (or even just one big one), theoverall area under the power spectrum density curve can be significantlyreduced by the presence of one or more large valleys, indicating a loweroverall transmission power for the UWB signal. Thus, a smoother powerspectrum density curve is preferable because that maximizes the areaunder the curve.

As it turns out, the more regular the transmitted pulse pattern (i.e.,the more predictable patterns are formed in the signal), the greater thenumber and size of the peaks and valleys in the curve. But if thetransmitted pulses appear random (i.e., they have no discernable patternof “1”s and “0”s), a smoother curve results. And since we want to get asmuch performance as we can from our assigned channel, it is desirable touse a signal that is smooth. This allows the signal to use a greateramount of power without violating the FCC's PSD constraints.

Alternate Types of Pulses

As noted above with respect to FIG. 1, in one embodiment of a UWBsystem, pulses (or chips) are short duration pulses formed using asingle basic pulse shape (e.g., a monopulse), with the interval betweenindividual pulses being uniform or variable. However, in alternateembodiments pulses can be formed in different ways.

In another preferred embodiment, portions of an oscillating carriersignal are used as pulses, e.g., three repetitions of the oscillatingsignal. These portions of the oscillating signal could be treated justas the pulses in FIG. 1, i.e., adjusted in amplitude or phase as shownin FIGS. 3A-3C. An example of a typical oscillating signal used to forma pulse stream in a preferred embodiment of the present invention isshown in FIG. 9. When these sorts of signals are used in a BPSK system,they can be referred to as n-cycle BPSK (e.g., the preferred embodimentof FIG. 9 is a three-cycle BPSK signal that uses three repetitions of abase oscillating signal).

As shown in FIG. 9, the carrier frequency of the oscillating signal(i.e., 1/T_(p-p)) in this embodiment is three times the chipping rate.In other words, the frequency of the waveform of the oscillating signalis three times the frequency of the pulses used by the network. Thisallows the network to take advantage of second order statistics that areunique to BPSK systems, and will allow improved acquisition.

Thus, although the embodiment of UWB using monopulses can be called acarrier-less radio system, it is also possible to use a carrier-basedsystem in which segments of the carrier are used to form pulses.

In particular, it means that it will be possible to recover the carrierfrequency off of BPSK sidebands by squaring the signal. And since thechipping frequency and the carrier frequency are related to each other,when you get the carrier frequency, you can easily calculate thechipping frequency.

FIGS. 10A-10C are graphs showing exemplary pulse streams for OOK, PPAM,and BPSK modulation schemes, respectively, using portions of anoscillating signal as pulses. In each case, they show a data sequence “10 1 0.”

As shown in FIG. 10A, OOK defines the data by the presence or absence ofa pulse. A “1” is indicated by a pulse, and a “0” is indicated by theabsence of a pulse. Thus, the bit stream “1 0 1 0” is indicated by thesequence of: a pulse, a blank where a pulse should be, a pulse, andanother blank. This embodiment has a_(k)ε{0,2}, i.e., data bits aretransmitted by either the presence or absence of a pulse at timet=t_(k).

As shown in FIG. 10B, PPAM defines the data by the amplitude of thepulse. A “1” is indicated by a large pulse, and a “0” is indicated by asmall pulse. Thus, the bit stream “1 0 1 0” is indicated by the sequenceof: a large pulse, a small pulse, a large pulse, and a small pulse. Thisembodiment uses strictly positive values for the two pulse weights, sothat a_(k)ε{α₀,α₁} where 0<α₀<α₁.

As shown in FIG. 10C, BPSK defines the data by the polarity of thepulse. A “1” is indicated by a non-inverted pulse, and a “0” isindicated by an inverted pulse. Thus, the bit stream “1 0 1 0” isindicated by the sequence of: a non-inverted pulse, an inverted pulse, anon-inverted pulse, and an inverted pulse. In this embodimenta_(k)ε{−1,+1}. This corresponds to transmitting either a non-inverted oran inverted pulse based on the value of the source bit.

Of course, this sort of pulse can also be used to form code words. FIG.11 is a timing diagram showing a five-pulse code word using repeatedoscillating signal pulses, according to a preferred embodiment of thepresent invention.

This embodiment has a code word that includes five binary pulses. Inthis case the code word period T_(cw) is five times the pulse periodT_(p) (i.e., the code words are transmitted at one-fifth the frequencyof the pulses).

In other words:T _(cw) =n*T _(p)  (16)

for an n-pulse code word. Thus, the pulse period T_(p) and number ofpulses n per code word determine the period of the code word T_(cw).

As shown in FIG. 11, a particular orientation of the five pulsescorresponds to a “1,” and the inverse of this orientation corresponds toa “0.” The particular choice of pulse orientation and arrangement withinthe code word is not critical, and can be varied as necessary. What isimportant is that the “1” and “0” code words are the inverse of eachother.

As noted above, although a five-pulse code word is shown for the sake ofsimplicity, 13-pulse or 24-pulse code words can be used in otherpreferred embodiments. In addition, in alternate embodiments, anysuitable number of pulses can be used to form a code word.

In one preferred embodiment the oscillating signal is a Gaussianmonopulse with a peak-to-peak time (T_(p-p)) of a fraction of ananosecond, a pulse period T_(p) of several nanoseconds, and a bandwidthof several gigahertz.

In alternate embodiments, different values can be used for T_(p-p) andT_(p). And in embodiments in which more or fewer repetitions are used todesignate a pulse, the relationship between T_(p-p) and T_(p) can alsovary.

Ternary Systems

Although the embodiments above show the use of binary values (i.e., 1and −1) for creating code words, it is also possible to use ternaryvalues (i.e., 1, 0, −1) to create the code words in alternate preferredembodiments. In this case, the code word will be made up not just of aseries of non-inverted and inverted pulses, but rather a series ofnon-inverted pulses, inverted pulses, and zeroed pulses. The zeroedpulses are preferably the absence of either a non-inverted pulse or aninverted pulse.

FIGS. 12A and 12B are timing diagrams showing a five-pulse code wordusing five ternary pulses, according to preferred embodiments of thepresent invention. FIG. 12A shows an embodiment using waveforms made ofmonopulses. FIG. 12B shows an embodiment using portions of a continuousoscillating waveform as pulses.

In each of these embodiments the code word period T_(cw) is five timesthe pulse period T_(p) (i.e., the code words are transmitted atone-fifth the frequency of the pulses).

In other words:T _(cw) =n*T _(p)  (17)

for an n-pulse code word. Thus, the pulse period T_(p) and number ofpulses n per code word determine the period of the code word T_(cw).

As shown in FIGS. 12A and 12B, a particular orientation of the fivepulses corresponds to a bit value of “1,” and the inverse of thisorientation corresponds to a bit value of “0.” The particular choice ofpulse orientation and arrangement within the code word is not critical,and can be varied as necessary. What is important is that the “1” and“0” code words are the inverse of each other.

However, unlike the embodiments shown above, these embodiments useternary pulses to form a code word. In comparison to a binary pulse,each ternary pulse can have a value of 1, 0, or −1. When performing theinverse function on ternary pulses, a “1” inverts to a “−1,” a “−1”inverts to a “1,” and a “0” remains a “0.”

In the particular embodiments shown, the code word is defined by thefive consecutive pulse values of 1 0 1 −1 −1, and the inverse code wordis defined by the five consecutive pulse values of −1 0 −1 1 1. In theembodiment of FIG. 12A, these values are imposed on monopulses. In theembodiment of FIG. 12B, these values are imposed on segments of acontinuous oscillating waveform.

Other than the fact that the pulse values are ternary rather thanbinary, they operate just as the code words described above with respectto FIGS. 7, 8, and 11.

For example, although FIGS. 12A and 12B each show a code word havingfive pulses, this number can be varied as needed. Alternate embodimentscan use any code word length that allows system requirements to be met.For example, as clock speeds increase, the number of pulses that can besent in a given time will increase and longer code word lengths may beused. In three preferred embodiments these code words are 12, 13, and 24ternary pulses in length.

Also, although the encoding in these embodiments is ternary, not binary,when multiple cycles of an oscillating signal are used to form pulses,this application will still refer to those pulses as n-cycle BPSKpulses.

Avoiding the UNII Band

Although the FCC will freely allow transmissions that do not violate thepower spectral density (PSD) limitations shown in FIG. 2, in someembodiments it is desirable to avoid certain areas of the bandwidthavailable for UWB transmissions. For example, in may be desirable tominimize overlap between UWB signals and the Unlicensed NationalInformation Infrastructure (UNII) band, which occupies the portion ofthe radio spectrum from 5.15 GHz to 5.825 GHz.

The UNII band is used by such devices as IEEE 802.11a devices, cordlesstelephones, HiperLAN2 devices, etc. By limiting overlap with this band,a UWB system can both reduce the chance that it will interfere withdevices that use the UNII band, and the chance that those devices willinterfere with transmissions by the UWB system.

Notches

One way to achieve this is by including a notch in the UWB transmissionscheme, the notch operating to minimize the amount of the UWB signalthat is transmitted within the UNII band. FIG. 13 is a graph of a UWBPSD using a notch according to a preferred embodiment of the presentinvention.

As shown in FIG. 13, the notch is placed within the UNII band such thatthe UWB signal's spectrum falls on either side of the UNII band. Thiscan be achieved by having a UWB system, whose carrier frequency iscentered in the UNII band, and which has a notch at the carrierfrequency. In this way, an RF spectral notch will fall right in the UNIIband. Most preferably, the notch is placed at a frequency slightly lowerthan the center of the UNII band to give maximum protection to the lowerand central portions of the UNII band. This is because most of the UNIIband devices operate in the lower or central portions of the band. Theupper range of the UNII band is primarily used for outdoorpoint-to-point links.

Multiple Bands

In addition, although UWB signals across a large spectrum can be formedas a single band, they can also be formed in two or more bands. Theseseparate bands each have a different power spectral density, and wouldbe limited to a subset of the total available bandwidth. By choosing theproper center frequencies, widths, and locations for these multiplebands, the power spectral density (PSD) for the UWB system can be moreeasily manipulated. For example, certain portions of the availablebandwidth can be avoided, or simply used with different PSD parametersthan other portions.

Each different overlapping device could then use a separate band.Although the transmissions of the overlapping devices would be sentwithin the same physical area, the interference each would experiencewith respect to each other would be limited by the separation of the twobands.

FIG. 14 is a graph of a UWB transmission scheme using two bandsaccording to a preferred embodiment of the present invention. As shownin FIG. 14, the high band has a center frequency of 8.208 GHz with a 3dB bandwidth of 2.736 GHz, and the low band has a center frequency of4.104 GHz with a 3 dB bandwidth of 1.368 GHz.

The high band in FIG. 14 preferably uses pulses that are made up ofthree iterations of an oscillating signal (e.g., a signal from a sinewave generator) having a frequency of 8.208 GHz (which can be called thehigh carrier frequency). Similarly, the low band preferably uses pulsesthat are made up of three iterations of an oscillating signal (e.g., asignal from a sine wave generator) having a frequency of 4.104 GHz(which can be called the low carrier frequency).

By choosing the frequency and placement of each band as they are, theUWB system is set such that both the low band has a −3 dB point at 4.788GHz, and the high band has a −3 dB point at 6.840 GHz. This gives a gapof 2.052 GHz between the two −3 dB points, just about in the middle ofthe UNII band. In this way, interference is minimized between the UWBtransmissions and any transmissions being made by other devices in theUNII band.

By having the high and low bands located where they are, and of thewidth they are, the UWB transmission system shown in FIG. 14 limits itsinterference with any other signals being transmitted in the UNII band,while also providing two separate transmission paths.

Alternate embodiments, however, could use a different number of bands,place the bands differently, and could use different types ofoscillating signals to form pulses.

Scaling Up Transmission Rate

In addition, one constant pressure in any transmission scheme, UWBincluded, is the desire to scale the transmission rate upward, (i.e.,send more data bits faster). For example, instead of sending a hundredmegabits, we want to send hundreds of megabits.

There are several ways that the data rate can be sped up. One way is tosend more pulses through within the same period of time. This wouldinvolve either reducing the width of the individual pulses or reducingthe spacing between adjacent pulses.

Another way is to use a smaller code word. As you drop pulses off of thecode word, the code word takes less time to transmit and thus allowsmore to be sent in a given time.

Yet another alternative is to use multiple code words to represent morethan just a binary bit of data. Rather than just using a code word andits inverse, you could use multiple different code words, eachrepresenting a different combination of multiple bits. For example, ifthe code word is five pulses long, there are thirty-two different waysthat inverted and non-inverted pulses can be combined to form a codeword. Counting for the fact that half of these will be the inverse ofothers, this allows for sixteen possible code words. This potentiallyallows up to five bits of data to be sent in the time it takes for fivepulses to be sent. The receiver can determine which bits of data havebeen sent by determining which combination of inverted and non-invertedpulses have been received as the code word.

More generally, this system allows for log₂(C) bits to be sent, where Cis the number of code words used. In the example above, if all 32 codewords were used, then log₂(32), or 5 bits could be sent.

However, there are some increased implementation costs with increasingthe number of different code words used. Because each receiver has tolook for each of the multiple code words used rather than just one, theymust each include additional circuitry to do so.

System for M-ary Bi-orthogonal Keying

FIGS. 15A and 15B are block diagrams of transmitter and receiver pairsaccording to preferred embodiments of the present invention. Thetransmitter and receiver pair 1500 a in FIG. 15A is used to producebinary code words. The transmitter and receiver pair 1500 b in FIG. 15Bis used to produce ternary code words.

As shown in FIG. 15A, the transmitter receiver pair 1500 a includes atransmitter 1510 a and a receiver 1520. The transmitter 1510 includes alookup table 1530, a pulse forming network (PFN) 1535 a, an adder 1540,and a transmitting antenna 1545. The receiver 1520 includes a receivingantenna 1550, a front end 1555, and a correlator 1560. As shown in FIG.15B, the transmitter receiver pair 1500 b includes a transmitter 1510 band a receiver 1520. The transmitter 1510 includes a lookup table 1530,a pulse forming network (PFN) 1535 b, an adder 1540, and a transmittingantenna 1545. The receiver 1520 includes a receiving antenna 1550, afront end 1555, and a correlator 1560. These two operate in the samemanner except for the operation of the PFN 1535 a, 1535 b in eachtransmitter 1510 a, 1510 b.

Transmitter

The lookup table 1530 receives a bit stream, breaks the bit stream upinto n-bit groups, and determines the proper code word associated withthat particular n-bit group. It then sequentially outputs a series of“1”s and “0”s corresponding to the proper code word. In this embodimentn can be any integer greater than 0. Although this preferred embodimentuses a lookup table 1530, alternate embodiments could use othercircuitry to perform this same function.

In the transmitter 1510 a or FIG. 15A, the PFN 1535 a receives thestring of “1”s and “0”s that define the code word from the lookup table1530 and outputs either a non-inverted or an inverted pulse in responseto each input value. In the preferred embodiment, the PFN 1535 areceives a clock signal CLK and a code word as inputs, and hasnon-inverted and inverted outputs. Whenever the clock CLK cycles, thePFN 1535 outputs either a non-inverted pulse at the non-inverted output,or an inverted pulse at the inverted output, depending upon the value ofthe individual portions of the code word.

In the transmitter 1510 b or FIG. 15B, the PFN 1535 b receives thestring of “1”s, “0”s, and “−1”s that define the code word from thelookup table 1530 and outputs either a non-inverted pulse, a zeroedpulse, or an inverted pulse in response to each input value. In thepreferred embodiment, the PFN 1535 b receives a clock signal CLK and acode word as inputs, and has non-inverted, zeroed, and inverted outputs.Whenever the clock CLK cycles, the PFN 1535 outputs either anon-inverted pulse at the non-inverted output, a zeroed pulse at thezeroed output, or an inverted pulse at the inverted output, dependingupon the value of the individual portions of the code word.

The pulses output from the PFN 1535 a, 1535 b can be any variety ofpulses. In one preferred embodiment, the pulses are individualmonopulses. In another preferred embodiment, the pulses are sectionsfrom an oscillating signal. Alternate embodiments could use otherpulses, if desired, however.

The adder 1540 then adds together the inverting and non-invertingoutputs (and zeroed output, if any), only one of which should be activeat a time, to provide a single output pulse. In the transmitter 1510 aof FIG. 15A, this output pulse will be either a positive (non-inverting)pulse or a negative (inverted) pulse, depending upon the value of thecurrent portion of the code word when the clock CLK cycles. In thetransmitter 1510 b of FIG. 15B, this output pulse will be either apositive (non-inverting) pulse, a zeroed pulse, or a negative (inverted)pulse, depending upon the value of the current portion of the code wordwhen the clock CLK cycles.

Alternate embodiments of the PFN 1535 could have a single output thatoutputs either a non-inverted or inverted pulse (or a non-inverted,zeroed, or inverted pulse) depending upon the value of the currentportion of the code word. In such embodiments there is no need for theadder 1540.

The output of the adder 1540 is then sent to the transmitting antenna1545, which transmits the pulses to the receiver 1520.

Receiver

The receiving antenna 1550 receives the pulses in a signal sent by thetransmitting antenna 1545 in the transmitter 1510.

The front end 1555 preferably performs necessary operations on thereceived signal to better allow the remainder of the receiver 1520 toproperly process it. This can include performing filtering andamplifying the signal.

The correlator 1560 receives a code word from the front end, determinewhat n-bit group corresponds to that code word (or inverse code word)and outputs the corresponding n-bit group. The correlator 1560 will haveto have as many different branches (called arms or fingers) to look forcode words as there are individual code words.

Single Code Word

In its simplest implementation, a single code word can be used to send abit stream from the transmitter 1510 to the receiver 1520 one bit ofdata at a time. The transmitter 1510 takes the bit stream, separates thestream into individual bits, chooses a code word/code word inverse basedon the bit to be transmitted, and then sends the chosen codeword/inverse to the receiver 1520.

The correlator 1560 in the receiver 1520 then needs to check eachincoming n-bits to see if they correspond to the code word or itsinverse. Since the correlator 1560 needs to look for only one code word,it needs only one arm (i.e., only one set of circuitry devoted tocorrelating a code word).

FIG. 16A is a block diagram of the correlator of FIGS. 15A and 15Bhaving one arm according to a preferred embodiment of the presentinvention. As shown in FIG. 16A, the correlator 1560 includes a mixer1610, an integrator 1615, and a decision circuit 1620. The mixer 1610receives the incoming signal and the code word. It mixes the code wordCW and a portion of the incoming signal equal in length to the code wordand outputs the result to the integrator 1615.

The integrator 1615 integrates the output of the mixer 1610 over thelength of the code word to produce a correlation result. Thiscorrelation result will be a large number if the code word is matched ora large negative number if the inverse of the code word is matched. Byexamining the correlation result, the decision circuit 1620 determineswhat data bit the code word corresponds to (i.e., a “1” or a “0”), andoutputs that data bit to other circuitry in the receiver 1520.

The single input stage (i.e., the mixer 1610) on the correlator 1560corresponds to the correlators 1560 one arm.

Multiple Code Words

However, as noted above, multiple code words can be used to sendmultiple bits at the same time. For example, to send two data bits at atime, two different code words should be used. Each code word thenrepresents two bits of data. In this case, if the bit stream to betransmitted were 0111001101001001, the transmitter would break it upinto two bit sections as so: (01)(11)(00)(11)(01)(00)(10)(01).

These two bits each represent four different choices, which correspondto the two code words and their inverses. One way this can beimplemented is to have the first bit determine which code word will beused, and have the second bit determine whether the chosen code word orits inverse should be used. This exemplary implementation is shown inTable 2. TABLE 2 Choice of Code Words by Bit Sequence Bit Sequence Codeword 00 First code word 11 Inverse of first code word 01 Second codeword 10 Inverse of second code word

The transmitter 1510 takes the bit stream to be sent to the receiver1520, separates the stream into 2-bit sections, chooses a code word/codeword inverse based on the two bits to be sent, and sends the chosen codeword/inverse to the receiver 1520.

The correlator 1560 in the receiver 1520 then needs to check eachincoming n-bits to see if they correspond to the first or second codeword or their inverses. Since the correlator 1560 needs to look for twocode words, it needs two arms (i.e., two sets of circuitry devoted tocorrelating a code word).

FIG. 16B is a block diagram of the correlator of FIGS. 15A and 15Bhaving two arms according to a preferred embodiment of the presentinvention. As shown in FIG. 16B, the correlator 1560 includes first andsecond mixers 1610 ₁ and 1610 ₂, first and second integrators 1615 ₁ and1615 ₂, and a decision circuit 1620. The mixers 1610 ₁ and 1610 ₂ eachreceive the incoming signal and one of the two code words CW₁ or CW₂.Each mixer 1610 ₁, 1610 ₂ mixes the respective code word and a portionof the incoming signal equal in length to the code words and outputs theresult to the first and second integrators 1615 ₁ and 1615 ₂,respectively.

The first and second integrators 1615 ₁ and 1615 ₂ integrate the outputof each of the mixers 1610 ₁ and 1610 ₂ over the length of the code wordto produce first and second correlation results, respectively. Eachcorrelation result will be a large number if the code word is matched, alarge negative number if the inverse of the code word is matched, or alower value if neither the code word nor its inverse is matched.

By examining the first and second correlation results, the decisioncircuit 1620 determines what two data bits the code word corresponds to(i.e., “00,” “01,” “10,” or “11”), and outputs those two data bits toother circuitry in the receiver 1520. This examination can be performed,for example, by setting a threshold for correlation values, above whicha correlation is considered successful, by comparing all of thecorrelation values and picking the largest result as a successfulcorrelation, or a combination of the two.

The double input stage (i.e., mixers 1610 ₁ and 1610 ₂) on thecorrelator 1560 corresponds to the two arms of the correlator 1560.

This can be extended beyond the use of two codes. In its general form, asystem that sends b bits of data must use a minimum of (k=2^((b-1)))code words. As the number of code words is increased, however, thereceiver still remains relatively simple, merely requiring k arms on itscorrelator 1560. The receiver 1520 still does not have to look forindividual pulses but instead looks for the k different code words inn-pulse increments.

The transmitter 1510 takes the bit stream to be sent to the receiver1520, separates the stream into n-bit sections (where n is greater than2), chooses a code word/code word inverse based on the n bits to besent, and sends the chosen code word/inverse to the receiver 1520.

The correlator 1560 in the receiver 1520 then needs to check eachincoming n-bits to see which of the first through k^(th) code words (orinverses) they correspond to. Since the correlator 1560 needs to lookfor k code words, it needs k arms (i.e., k sets of circuitry devoted tocorrelating a code word).

FIG. 16C is a block diagram of the correlator of FIGS. 15A and 15Bhaving more than two arms according to a preferred embodiment of thepresent invention. As shown in FIG. 16C, the correlator 1560 includesfirst through k^(th) mixers 1610 ₁ to 1610 _(k), first through k^(th)integrators 1615 ₁ to 1615 _(k), and a decision circuit 1620. The mixers1610 ₁ to 1610 _(k) each receive the incoming signal and one of the kcode words CW₁ to CW_(k). Each mixer 1610 ₁, . . . , 1610 _(k) mixes therespective code word and a portion of the incoming signal equal inlength to the code words and outputs the result to the first throughk^(th) integrators 1615 ₁ to 1615 _(k), respectively.

The first through k^(th) integrators 1615 ₁ and 1615 _(k) integrate theoutput of each of the mixers 1610 ₁ to 1610 _(k) over the length of thecode word to produce first through k^(th) correlation results,respectively. Each correlation result will be a large number if therespective code word is matched, a large negative number if the inverseof the respective code word is matched, or a lower value if neither therespective code word nor its inverse is matched.

By examining the first through k^(th) correlation results, the decisioncircuit 1620 determines what b data bits the code word corresponds to,and outputs those b data bits to other circuitry in the receiver 1520.This examination can be performed, for example, by setting a thresholdfor correlation values, above which a correlation is consideredsuccessful, by comparing all of the correlation values and picking thelargest result as a successful correlation, or a combination of the two.

The multiple input stage (i.e., mixers 1610 ₁ through 1610 _(k)) on thecorrelator 1560 corresponds to the k arms of the correlator 1560.

In each embodiment the correlator 1560 preferably looks for the codewords rather than individual pulses. If the correlator 1560 looked foreach individual pulse, it would have to make a lot more decisions, sinceit would check a larger number of individual pulses. The complexity ofthe system can be significantly reduced by having the correlator 1560look instead for a long sequence of pulses.

When multiple code words are used, the choice of how the code words areformed becomes important. Although for any given code word size n, thereare 2^((n-1)) possible code words to choose from, it turns out that somecode words are better than others. One reason for this is the operationof the correlator 1560 in the receiver 1520.

When multiple code words are used, the correlator 1560 has multiple arms(i.e., multiple mixers 1610 ₁ to 1610 _(k)). The decision circuit 1620has to take the first through k^(th) correlation results from thesemixers 1610 ₁ to 1610 _(k) and compare them to determine which code word(or inverse) was received. By choosing code words that are orthogonal toeach other, the function performed by the decision circuit 1620 can begreatly simplified.

As noted above, when a particular code word (or its inverse) sent fromthe transmitter 1510 is received by the i^(th) mixer 1610 _(i) and ismixed with the respective code word CW_(i), the i^(th) mixer 1610 _(i)will output a large positive number if the received code word matchesthe i^(th) code word, and will output a large negative number if thereceived code word matches the inverse of the i^(th) code word.

If the code words are not orthogonal to each other, then if the receivedcode word does not match the i^(th) code word CW_(i), the i^(th) mixerwill output a non-zero value somewhere between the large negative numberand the large positive number. However, if the code words are orthogonalto each other, then if the received code word does not match the i^(th)code word CW_(i), the i^(th) mixer will output a value of zero.

One way the decision circuit 1620 can decide which correlation value isthe correct one (i.e., which indicates the proper code word) is that itexamines all of the outputs of the receiver mixers 1610 ₁ to 1610 _(k)and looks for the one that has the largest absolute value. Ideally, theother receiver multipliers should have an output close to zero.

This indicates which of the code words has been received, but notwhether it was inverted or not inverted. For this the decision circuit1620 looks at the sign of the chosen correlation value. If it ispositive, then the non-inverted code word has been received; if it isnegative, then the inverted code word has been received. By usingorthogonal code words, the system allows the receiver to make a mucheasier comparison of correlation values, since when the code words areorthogonal, one correlation value will be a large positive or negativenumber (i.e., the one corresponding the received code word), and all ofthe other correlation values will be zero (or close to zero, accountingfor the presence of noise in the system). Thus, when the absolute valueof the output of the correlator corresponding to the received code wordis at a maximum, the output of the other correlators will be zero.

Thankfully, orthogonal code sets are not difficult to find. For example,if you have a code word of twelve pulses, (i.e., a length twelve codeword), you would have 2¹² or 4096 possible code words. From this set of4096 possible code words there is at least one set of twelve codes thatare mutually orthogonal, as shown in Table 3. Other orthogonal subsetsalso exist. TABLE 3 Orthogonal Code Word Set for Length 12 Code WordCode Word B₀ B₁ B₂ B₃ B₄ B₅ B₆ B₇ B₈ B₉ B₁₀ B₁₁ 1 −1 1 1 1 1 1 −1 −1 −11 1 −1 2 −1 −1 −1 −1 −1 −1 −1 −1 −1 −1 1 −1 3 −1 −1 −1 −1 1 1 −1 −1 1 1−1 1 4 −1 −1 −1 1 −1 1 1 1 −1 1 −1 −1 5 −1 −1 1 −1 1 −1 1 1 −1 1 1 1 6−1 −1 1 1 −1 1 −1 1 1 −1 1 1 7 −1 −1 1 1 1 −1 1 −1 1 −1 −1 −1 8 −1 1 −1−1 1 1 1 1 1 −1 1 −1 9 −1 1 −1 1 −1 −1 1 −1 1 1 1 1 10 −1 1 −1 1 1 −1 −11 −1 −1 −1 1 11 −1 1 1 −1 −1 −1 −1 1 1 1 −1 −1 12 −1 1 1 −1 −1 1 1 −1 −1−1 −1 1

Tables 4A-4D show examples of sets of orthogonal 24-bit code words forup to four overlapping networks. Table 4A shows the code words used forthe first network; Table 4B shows the code words used for the secondnetwork; Table 4C shows the code words used for the third network; andTable 4D shows the code words used for the fourth network. TABLE 4AOrthogonal Code Word Set for Length 24 Code Word - Network #1 Code WordAssociated 24-bit pattern 0000 −1 1 −1 −1 1 −1 −1 1 −1 0 −1 0 −1 −1 1 11 −1 1 1 1 −1 −1 −1 0001 0 −1 −1 0 1 −1 −1 1 −1 −1 1 1 1 1 −1 −1 1 −1 1−1 1 1 1 1 0010 −1 −1 −1 −1 1 −1 1 −1 1 −1 −1 1 −1 −1 1 −1 −1 1 1 0 −1 01 1 0011 0 −1 1 1 1 −1 −1 −1 −1 −1 −1 −1 1 −1 1 −1 0 1 −1 1 1 −1 −1 10100 −1 0 1 −1 −1 −1 1 1 0 1 1 1 1 −1 1 −1 1 1 1 −1 1 −1 −1 1 0101 −1 0−1 1 −1 1 −1 −1 0 1 1 1 1 −1 1 1 −1 −1 −1 1 1 −1 1 1 0110 −1 −1 −1 −1 −1−1 1 1 1 0 −1 −1 1 1 −1 1 −1 1 −1 1 1 −1 0 1 0111 −1 1 −1 −1 −1 1 −1 −10 −1 1 −1 −1 1 −1 0 1 1 1 1 −1 −1 −1 1

TABLE 4B Orthogonal Code Word Set for Length 24 Code Word - Network #2Code Word Associated 24-bit pattern 0000 −1 −1 1 0 1 1 1 −1 −1 1 −1 1 1−1 1 0 1 −1 −1 −1 1 −1 −1 −1 0001 −1 −1 −1 1 −1 −1 −1 1 0 1 −1 1 1 −1 1−1 −1 1 1 1 0 1 −1 −1 0010 −1 1 −1 1 1 −1 1 0 1 1 1 −1 −1 1 1 −1 1 1 1−1 −1 −1 0 −1 0011 0 −1 1 1 1 1 −1 −1 1 1 1 −1 1 1 −1 1 1 1 −1 1 −1 0 −1−1 0100 −1 1 −1 1 −1 −1 −1 −1 −1 −1 −1 1 1 1 −1 −1 1 1 −1 0 1 −1 0 10101 −1 1 −1 −1 1 0 −1 −1 1 1 −1 −1 0 1 1 1 −1 −1 −1 −1 −1 1 −1 1 0110−1 0 1 −1 −1 −1 1 −1 1 −1 1 1 1 1 −1 −1 −1 −1 1 −1 0 1 −1 −1 0111 −1 −1−1 −1 −1 −1 1 1 1 0 −1 1 −1 1 −1 1 1 −1 −1 1 −1 0 1 −1

TABLE 4C Orthogonal Code Word Set for Length 24 Code Word - Network #3Code Word Associated 24-bit pattern 0000 −1 1 −1 1 −1 −1 0 1 −1 −1 −1 1−1 −1 1 0 −1 −1 −1 −1 1 1 1 1 0001 −1 −1 1 1 −1 −1 −1 −1 −1 −1 1 1 0 1−1 1 1 −1 1 −1 0 −1 1 −1 0010 −1 −1 −1 1 1 1 −1 −1 −1 1 −1 −1 −1 1 −1 −11 −1 1 0 1 1 0 1 0011 −1 −1 1 −1 −1 1 1 1 −1 −1 1 −1 −1 −1 −1 0 1 1 −1 1−1 1 0 1 0100 −1 −1 −1 1 −1 1 −1 1 0 −1 −1 −1 1 1 1 1 −1 1 1 −1 0 1 −1−1 0101 −1 −1 −1 0 −1 −1 −1 −1 1 1 1 0 1 −1 −1 1 −1 1 −1 1 1 −1 −1 10110 −1 1 −1 1 −1 1 1 0 1 1 1 0 −1 1 1 −1 1 1 −1 −1 −1 −1 1 1 0111 −1 10 −1 1 −1 1 −1 −1 −1 1 −1 −1 0 1 −1 −1 1 1 1 1 −1 −1 −1

TABLE 4D Orthogonal Code Word Set for Length 24 Code Word - Network #4Code Word Associated 24-bit pattern 0000 −1 −1 1 1 1 −1 −1 −1 −1 −1 −1 0−1 1 −1 1 −1 1 1 −1 1 1 −1 0 0001 −1 −1 −1 1 −1 1 1 1 1 −1 1 1 −1 1 1 −1−1 1 1 1 0 0 −1 1 0010 −1 1 −1 1 1 1 1 0 −1 −1 −1 −1 1 −1 0 −1 −1 1 1 −1−1 1 1 −1 0011 0 −1 −1 −1 −1 −1 −1 1 1 0 −1 1 1 −1 1 −1 −1 1 1 −1 1 −1 1−1 0100 −1 −1 1 1 −1 −1 1 0 −1 1 1 1 1 −1 1 −1 1 −1 0 −1 1 1 1 1 0101 −1−1 1 −1 −1 1 −1 −1 0 −1 1 −1 1 1 −1 −1 −1 1 −1 0 −1 1 1 1 0110 −1 1 0 1−1 −1 −1 1 1 −1 0 −1 1 −1 −1 1 −1 −1 1 1 1 1 1 1 0111 −1 −1 −1 −1 1 −1 10 −1 1 −1 1 1 1 0 1 −1 −1 1 1 −1 −1 1 1

Furthermore, since the system uses not only the chosen orthogonal codewords, but also their inverses, there are actually twice as manypossible codes to choose from. (Hence the term “bi-orthogonal.”) In theexample above where twelve code words were chosen, there will actuallybe twenty-four possible codes—twelve code words and their inverses.

Thus, if you use a length twelve code word and pick eight of the abovetwelve orthogonal code words (sixteen total possible values, countinginverses), you can send four bits of data at a time. (16=2⁴)

Stated more generally, if b is the number of data bits you want to send,you need to find a group of 2^((b-1)) mutually orthogonal code words.Adding their inverses, this will give you 2^(b) total codes, which issufficient to send b bits of data.

In alternate embodiments you could design an orthogonal system thatdoesn't use inverses. However, then you would need 2^(b) mutuallyorthogonal codes to send b bits of data, instead of 2^((b-1)) mutuallyorthogonal codes. Thus, you would need twice as many codes if you don'tuse their inverses.

In contrast, by using the inverses you can use half as many codesbecause of the inverses make up the other half. Using inverses alsogives slightly better performance and the potential for having nospectral lines.

Within the set of possible code words (2^(p) combinations of pulses,where p is the number of pulses in the code word), there will benumerous different subsets that are mutually orthogonal. Any of thesesubsets can be used, provided they have enough elements to pass therequired number of data bits. Not surprisingly, the larger the number ofpulses in the code word and the smaller the number of data bits sent,the easier it will be to find a useful subset of mutually orthogonalcodes.

In alternate embodiments it is also possible to use code sets that arenearly orthogonal. Nearly orthogonal refers to code word sets in whichthe correlation between different code words from the set gives resultsthat are greater than zero, but are smaller than an acceptablethreshold.

This threshold can be determined based on the ability of the correlatorto differentiate between a correct code word correlation and anincorrect code word correlation. One possible threshold is a randomcorrelation threshold, which corresponds to the expected correlationvalue that random noise would give. In other words, using thisthreshold, a set of codes would be considered nearly orthogonal if thecorrelation between all combinations of different code words in the setgive results that are better than the expected correlation between anyof the code words and a set of random bits equal in length to the codeword. However, alternate embodiments could use a different threshold.

By allowing the use of nearly orthogonal code sets for the code words,the system greatly increases the available choices for code word sets.

Thus, by using multiple code words, it is possible to send data at agreater speed without sending individual pulses any quicker. Forexample, if four code words are used (allowing three data bits to besent at a time), a system that would operated at 100 MBPS with one codeword could operate at 300 MBPS without significantly increasing thecomplexity of the transmission circuitry at all. The price that willhave to be paid is to put multiple arms into the correlators in eachreceiver to look for all these code words at the same time.

Additional Advantages of Having Multiple Correlators

One added advantage of using more than one arm in the correlator 1560(as shown for example, in FIGS, 16B and 16C) is that other functions ofthe receiver 1520, 1720 can also make use of these features, Forexample, during signal acquisition, multiple arms can be used not tolook for multiple code words, but to find a single code more quickly,speeding up the acquisition process. Likewise, in a clutteredenvironment where a signal bounces off walls or other obstructions andas a result the receiver 1520, 1720 receives multiple copies of the samesignal (often referred to as multipath), the multiple arms can trackdifferent time-displaced versions of the same signal to determine whichprovides the strongest signal. By tracking the same code with twodifferent arms, the receiver 1520, 1720 can get better performance.

Thus, the multiple arms in the correlator (i.e., the mixers 1610 _(i))can be used for fast acquisition, multipath reception, or multiple codewords. When the signal is initially being acquired, the multiple armsare used to speed up acquisition. If, after the signal is acquired, thesignal quality is low due to the multipath effect, the multiple arms canthen be used to track different time-shifted versions of the samesignal. If, however, signal quality is good, then the multiple arms canbe used to accommodate multiple code words and thereby speed up datatransmission.

Thus, the multiple arm structure is configurable into a variety of usesthat can improve system speed or performance. And if none of these usesare required, the additional arms of the correlator can be turned off tosave power.

Benefits Regarding Analog-to-Digital Converters

Another advantage to this design is that by using multiple arms in eachcorrelator 1560, the receiver need not increase the complexity of theanalog-to-digital converters (ADCs) contained in the correlator 1560 asthe data rate is increased.

Since the decision circuit 1620 in the correlator 1560 receives ananalog correlation signal from each mixer 1610 _(i), and outputs adigital data signal to another portion of the receiver 1520, it willhave to have at least one ADC to convert the received analog signal to adigital signal.

If the receiver 1520 used a single arm in its correlator 1560 (i.e., asingle mixer 1610 checking for correlation with a single code word), theADC in the decision circuit 1620 would need to operate at the full datatransmission speed. For example, if the receiver 1520 was operating at400 MBPS, the ADC would also have to operate at 400 MBPS—making 400million decisions every second.

However, if the receiver 1520 used four arms in its correlator 1560(i.e., four mixers 1610 ₁ to 1610 ₄ to check for four separate codewords CW₁ to CW₄ at the same time), then it would require four ADCs inthe decision circuit 1620, each operating at 100 MBPS—making only 100million decisions every second. This makes for a simpler design, sinceit is easier to design lower speed ADCs. As the speed of ADCs increasethey have to have a higher performance, be of higher quality, have lessdistortion, etc. than a lower speed ADC. This is usually more expensive,may take up more chip area, and may require a more expensive processingtechnology to create than a larger number of slower ADCs. Generally itis better to use multiple slow ADCs than one fast one.

In alternate embodiments it is also possible to make all of thenecessary correlation decisions before any analog-to-digital conversion,thus using only one slow ADC.

Scalability

Furthermore, the concept of multiple code words is scalable, with theonly limit being the number of correlators required in the receivers. Ifyou want to send b bits at a time, then you need 2^((b-1)) code wordsand therefore 2^((b-1)) arms in the correlators 1560 of each receiver1520.

Currently the number of bits b sent at a time is preferably between 2and 5, i.e., 4 to 16 arms in each correlator 1560. However, assemiconductor fabrication technology advances, the number of workablecorrelators in a receiver will expand and the number of usable codeswill similarly increase.

Power Efficiency

Another advantage of this design is that it has improved powerefficiency. By using M-ary bi-orthogonal keying (MBOK), a transmissionwill suffer fewer errors for the same amount of noise that's present inthe system as compared to BPSK or PPM. This results in a coding gain,i.e., a more power efficient modulation scheme. In other words, for thesame amount of errors at the receiver 1520, the transmitter 1510 woulduse less transmission power. Equivalently for the same amount oftransmission power, the receiver 1520 would suffer fewer errors.

In a system that has a set number of allowable errors, e.g., 10⁻³ BER,this means that the transmitter 1510 can either transmit at a lowerpower to achieve that BER, or the signal could be sent over a longerdistance before it reached its maximum BER because of the increasedpower efficiency. In other words, the system is more robust to errorsallowing better range and better performance.

Alternative Solutions

As noted above, two alternative solutions for increasing datatransmission rate is to either use shorter codes or to push the pulsescloser together. However, both of these approaches have very diminishingreturns. This is because without using a solution such as multiple codewords, if you want to make the bit rate four times as fast, then youhave to send four times as many pulses. That requires you to have thesystem clocks running four times as fast, which takes a lot more power,and power is of paramount importance for any portable device.

The alternative of pushing the pulses closer together also runs into theproblem noted above that there is a physical limitation on how closethey can be pushed together (i.e., the width of the pulse plus theminimum separation between pulses). Once you reach that point, the speedof the device cannot be increased in that way.

The alternative of picking smaller code words runs into problems of thePSD of the transmission. The shorter the code words, the more repetitivethe overall data transmission will be, and the higher the peaks in thePSD of the transmission. Given the restrictions imposed by the FCC (SeeFIG. 2), it's preferable to keep these peaks as low as possible tomaximize the transmission power of the signal. Therefore, as the codewords are made shorter, the PSD of the transmitted signal deteriorates.

In addition, as smaller code words are chosen, it will be harder to findgroups of mutually orthogonal codes within the set of possible codewords. At some point, if the code words are reduced in length enough, itwill be impossible to find the requisite number of mutually orthogonalcode words.

Pseudo-Random Scrambling

As shown in FIG. 2, the FCC has limited the power spectral density (PSD)of UWB signals. As noted above, it is preferable to keep the PSD of atransmitted signal as smooth as possible to maximize the allowabletransmission power for the signals.

One way to smooth out the PSD in a code word implementation is to usevery long code words. The longer the code words, the more they will looklike random transmissions, and the smoother the PSD curve. However, ifthe code words are made longer than they need to be, availabletransmission time will be wasted, reducing the maximum data rate allowedby the system.

This is made even more difficult in an MBOK system by the requirementthat the code words chosen must be orthogonal to each other. The codewords must be chosen first for their orthogonality, and only then cantheir average spectrum properties be considered. Given the limitednumber of mutually orthogonal sets of codes, it can be extremely hard tochoose a set of code words that do not result in any sharp peaks in thePSD curve.

One solution to this problem is to multiply the data signal by apseudo-random sequence to whiten it before transmission. Thepseudo-random sequence is a sequence of +1 and −1 values that ispredictable and preferably longer than a code word, but looks random(i.e., has no discernable pattern). By multiplying the transmitted codewords with this pseudo-random pattern they will be scrambled such thatthey appear random, but will be scrambled in a way that can beunscrambled at the receiver. This will allow bi-orthogonal keying to beused, but provide acceptable spectrum properties.

The pseudo-random sequence should be either known to both transmitterand receiver, or there should be a deterministic way of producing thispattern, e.g., performing a function that starts from a known sequence,that is known by both the transmitter and the receiver. Thus both thetransmitter and the receiver will be able to produce the pseudo-randomsequence and the scrambling can be both performed and undone. This willmake the scrambling completely transparent to the rest of the system.

Specifically, by multiplying the transmitted code words sequentially bythe pseudo-random series of +1 and −1 values, the transmitter will takea string of code words (or inverses) and scramble them into a pseudorandom pattern. This will flatten out the spectrum of the transmittedsignal, reducing the number of sharp peaks in its PSD. This eliminatesthe need to worry about the spectrum properties of the chosen codewords, since the pseudo-random scrambling eliminates any short-termregularity in those codes, essentially whitening the transmission. Thenthe receiver reverses the process and obtains the encoded data. Thus,pseudo-random scrambling enables the system use the advantageousproperties of bi-orthogonal keying, but with superior power efficiency.

Pseudo-Random Scrambling Circuitry

FIGS. 17A and 17B are block diagrams of transmitter and receiver pairsusing pseudo-random scrambling according to preferred embodiments of thepresent invention. The transmitter and receiver pair 1700 a in FIG. 17Ais used to produce binary code words. The transmitter and receiver pair1700 b in FIG. 17B is used to produce ternary code words.

As shown in FIG. 17A, the transmitter receiver pair 1700 a includes atransmitter 1710 a and a receiver 1720. The transmitter 1710 a includesa lookup table 1530, a pulse forming network (PFN) 1535 a, an adder1540, a transmitting antenna 1545, a transmitter mixer 1770, atransmitter pseudo-random sequence generator 1775, and a transmitterswitch 1778. The receiver 1720 includes a receiving antenna 1550, afront end 1555, a correlator 1560, a receiver mixer 1780, a receiverpseudo-random sequence generator 1785, a receiver switch 1788, and anacquisition circuit 1790. As shown in FIG. 17B, the transmitter receiverpair 1700 b includes a transmitter 1710 b and a receiver 1720. Thetransmitter 1710 b includes a lookup table 1530, a pulse forming network(PFN) 1535 b, an adder 1540, a transmitting antenna 1545, a transmittermixer 1770, a transmitter pseudo-random sequence generator 1775, and atransmitter switch 1778. The receiver 1720 includes a receiving antenna1550, a front end 1555, a correlator 1560, a receiver mixer 1780, areceiver pseudo-random sequence generator 1785, a receiver switch 1788,and an acquisition circuit 1790. These two operate in the same mannerexcept for the operation of the PFN 1535 a, 1535 b in each transmitter1710 a,1710 b.

The elements in FIGS. 17A and 17B that are the same as FIGS. 15A and 15Boperate in the same or similar manner and so their description will notbe repeated.

The Transmitter

The transmitter pseudo-random sequence generator 1775 operates togenerate a long pseudo-random sequence. In the preferred embodiment, thetransmitter pseudo-random sequence generator 1775 is a shift registerthat has the pseudo-random sequence entered into it. More specifically,in the preferred embodiment the transmitter pseudo-random sequencegenerator 1775 is a liner feedback shift register.

In this embodiment, the transmitter pseudo-random sequence generator1775 has a number of storage locations that contain the long,pseudo-random sequence (i.e., a long string of pseudo-random +1 and −1values). In a preferred embodiment the transmitter pseudo-randomsequence generator 1775 contains between fifteen and thirty entries,though it may have more or fewer in alternate embodiments.

The pseudo-random sequence is shifted through the pseudo-random sequencegenerator 1775 each time the lookup table 1530 outputs a pulse, and thetop entry is multiplied with the output of the lookup table 1530 at thetransmitter mixer 1770. (Preferably this is an XOR operation.) Thetransmitter pseudo-random sequence generator 1775 (i.e., the shiftregister) may also have one or more feedback taps that help make thesequence appear more random. These feedback taps allow a bit from withinthe shift register to combined with the ending bit being cycled back tothe beginning of the shift register. This feedback can help make thepseudo-random sequence of “1”s and “0”s appear more random withoutbecoming any less predictable for the receiver 1720.

In alternate embodiments a different circuit could be used as thetransmitter pseudo-random sequence generator 1775. For example, thetransmitter pseudo-random sequence generator 1775 could be a circuitthat performs a known function that outputs a pseudo-random sequence.

It is preferable that the transmitter 1710 also have the capability tocontrol when the pseudo-random sequence is provided to the transmittermixer 1770. To this end, the transmitter switch 1778 is provided. Whenthe transmitter switch 1778 is open, the transmitter mixer 1770 passesthe pulses from the lookup table 1530 to the PFN 1535 a, 1535 bunchanged (i.e., it multiplies them by a constant value of +1). When thetransmitter switch 1778 is closed, however, the pulse stream from thelookup table 1530 is multiplied by the pseudo-random sequence from thetransmitter pseudo-random sequence generator 1775. One reason for thisis that some transmissions need not be sent with a pseudo random elementadded.

However, when the transmitter switch 1778 is closed, the code wordsoutput by the lookup table 1530 will be whitened by the pseudo-randomsequence stored in the transmitter pseudo-random sequence generator 1775before they are transmitted by the transmitting antenna 1545.

Although FIGS. 17A and 17B show that the pseudo-random sequence isintroduced between the lookup table 1530 and the PFN 1535 a, 1535 b, inalternate embodiments it could also be introduced after the PFN 1535 a,1535 b. However, for implementation reasons, it is preferable that thepseudo-random element be added before the pulses are generated.

One time when the transmitter 1710 a, 1710 b and receiver 1720 may notuse pseudo-random scrambling is during signal acquisition. When a datapacket is first sent from a transmitter to a receiver, a certain periodof time will be spent synchronizing the two devices up.

FIG. 18 is a block diagram of a data packet according to a preferredembodiment of the present invention. As shown in FIG. 18, the packet1800 includes a preamble 1810, a header 1820, and data 1830. Eachportion of the packet is made up of a series of pulses representing thebits of data in that portion of the packet 1800.

In the preamble 1810, the transmitter 1510 sends a known sequence ofsignals (e.g., a pattern of one particular code word and its inverse).The receiver 1520 listens for this known sequence in order to properlylock onto the signal from the transmitter 1510. No substantive data issent in the preamble 1810 since the receiver 1520 is still getting itstiming synchronized with that of the transmitter 1510. The header 1820includes information about the intended recipient of the packet 1800 andother identifying information. The data 1830 includes the substantivedata being transmitted by the packet 1800.

Preferably, the preamble 1810 is not scrambled with a pseudo-randomsequence, but the header 1820 and the data 1830 are scrambled. This isbecause for proper synchronization to occur, the receiver 1520 must knowwhat the preamble sent by the transmitter 1510 looks like. This wouldnot be possible if the preamble 1810 were scrambled and the receiver didnot yet have information (contained in the preamble 1810) necessary todescramble it.

Thus, in this embodiment the transmitting switch 1778 is kept open whilethe preamble 1810 is being sent, and is closed when the header 1820 anddata 1830 are being sent. Likewise, the receiving switch 1788 will bekept open while the preamble 1810 is being received, and is closed whenthe header 1820 and data 1830 are being received.

The Receiver

The receiver pseudo-random sequence generator 1785 generates the samelong pseudo-random sequence contained in the transmitter pseudo-randomsequence generator 1775. In the preferred embodiment, the receiverpseudo-random sequence generator 1785 is also a shift register, morepreferably a linear feedback shift register. The pseudo-random sequenceis preferably generated and entered into the receiver pseudo-randomsequence generator 1785. The receiver shift register has a number ofstorage locations that contain the long, pseudo-random sequence (i.e., along string of pseudo-random +1 and −1 values). In a preferredembodiment the receiver shift register contains between fifteen andthirty entries, though it may have more or fewer in alternateembodiments.

The pseudo-random sequence is shifted through the receiver pseudo-randomsequence generator 1785 each time a pulse is received from the front end1555, and the top entry is multiplied with the output of the front end1555 at the receiver mixer 1780. (Preferably this is performed with anXOR function.) The receiver shift register may also have one or morefeedback taps that help make the sequence appear more random. Thesefeedback taps have a bit from within the shift register combined withthe ending bit being cycled back to the beginning of the shift register.As a result the receiver pseudo-random sequence generator 1785 outputs apseudo-random sequence of “1”s and “0”s to the receiver mixer 1780.

The operation of the receiver pseudo-random sequence generator 1785 issynchronized with the operation of the transmitter pseudo-randomsequence generator 1775 such that they both output the samepseudo-random sequence. Thus, although the sequence appears random, itis also deterministic.

As with the transmitter pseudo-random sequence generator 1775, inalternate embodiments a different circuit could be used as the receiverpseudo-random sequence generator 1785. For example, the receiverpseudo-random sequence generator 1785 could be a circuit that performs aknown function that outputs a pseudo-random sequence.

It is preferable that the receiver 1720 also have the capability tocontrol when the pseudo-random sequence is provided to the receivermixer 1780. To this end, the receiver switch 1788 is provided. When thereceiver switch 1788 is open, the receiver mixer 1780 passes the pulsesfrom the front end 1555 to the correlator 1560 unchanged (i.e., itmultiplies them by a constant value of +1). When the receiver switch1788 is closed, however, the pulse stream from the front end 1555 ismultiplied by the pseudo-random sequence from the receiver pseudo-randomsequence generator 1785 before it is sent to the correlator 1560. Onereason for this is that some signals have been sent with no pseudorandom element added (e.g., the preamble 1810).

Thus, when the receiver switch 1778 is closed, the code words receivedat the receiver antenna 1550 and processed by the front end 1555 will bedescrambled by the pseudo-random sequence output by the receiverpseudo-random sequence generator 1785 before they are correlated by thecorrelator 1560.

Although FIGS. 17A and 17B show that the pseudo-random sequence isintroduced between the front end 1555 and the correlator 1560, inalternate embodiments it could also be introduced before the front end1555. However, for implementation reasons, it is preferable that thepseudo-random element be added after the received pulses are amplified.

As noted above, when the signal coming into the receiver 1720 isscrambled (e.g., during the header 1820 and the data 1830 portions ofpacket 1800), the receiver 1720 will have to reverse thepseudo-randomization of the pulses in the signal.

The receiver 1720 accomplishes this based on information received duringthe unscrambled preamble 1810. During the preamble 1810 of a givenpacket 1800, the acquisition circuit 1790 in the receiver 1720 performsseveral functions. First, it locks onto the timing of the incomingsignal by identifying a known sequence of pulses contained in thepreamble 1810 (e.g., one of the code words and its inverse). Based onthis timing information, the acquisition circuit 1790 then determineswhere the boundaries between code words are. Finally, it determines whenthe preamble 1810 ends and the header 1820 begins, and instructs thereceiver switch 1788 to close so that the incoming pseudo-random streamof pulses can be descrambled, and instruct the pseudo-random sequencegenerator 1785 to begin. The receiver switch 1788 will then be openedafter the last of the data is received, to await a new packet.

As noted above, the receiver pseudo-random sequence generator 1785outputs the same pseudo-random stream of +1 and −1 values as thetransmitter pseudo-random sequence generator 1775. Since the receiverpseudo-random sequence generator 1785 starts at the same position as thetransmitter pseudo-random sequence generator 1775, the operation of thereceiver pseudo-random sequence generator 1785 will undo the scramblingcaused by the transmitter pseudo-random sequence generator 1775.

After being multiplied by the corresponding value from the pseudo-randomstream in the receiver mixer 1780, the data stream will be back to theway it was before being multiplied by the corresponding value from thepseudo-random stream in the transmitter multiplier 1770. If the valuefrom the pseudo-random stream was +1, then the pulse will be unchangedby either operation; if the value from the pseudo-random stream was −1,then the pulse will be inverted by the first operation and returned tonormal by the second operation.

The conceptual block diagram of FIG. 17A could be used with three-cycleBPSK (i.e., using a pulse of three cycles of an oscillating signal). Thegeneration of the three-cycle wavelet in the transmitter and thedemodulation of the three-cycle wavelet in the receiver are performed inthe PFN 1535 a, and the correlator 1560, respectively.

A random overlay cover code using the wavelets of +1 and −1 for the codecould also be used for ternary M-BOK code words. In preferredembodiments the cover code will always be binary (i.e., it will usevalues of ±1), and not ternary, regardless of the underlying code words.

Operation of the Transmitter and Receiver

FIGS. 19 and 20 are flow charts describing the operation of thetransmitter and receiver, respectively, according to a preferredembodiment of the present invention.

As shown in the flow chart of FIG. 19, the process begins when thetransmitter 1510 a, 1510 b, 1710 a, 1710 b (See FIGS. 15A, 15B, 17A, and17B) receives a bit stream. (Step 1910) The transmitter 1510 a, 1510 b,1710 a, 1710 b then breaks b bits off of the bit stream (Step 1920) anddetermines a code word (or code word inverse) that corresponds to thecurrent b-bit segment from the bit stream. (Step 1930)

If pseudo-random scrambling is to be used on the current code word, thetransmitter 1510 a, 1510 b, 1710 a, 1710 b then multiplies the code wordby a predictable pseudo-random sequence. (Step 1940) In embodimentswhere no pseudo-random scrambling is used (e.g., the embodiments shownin FIGS. 15A and 15B), or in portions of a packet where no pseudo-randomscrambling is used, the transmitter 1510 a, 1510 b, 1710 a, 1710 b canomit this step.

The transmitter 1510 a,1510 b, 1710 a, 1710 b then transmits the codeword (scrambled or unscrambled) to the receiver 1520, 1720 (See FIGS.15A, 15B, 17A, and 17B). (Step 1950)

Once this is done, the transmitter 1510 a, 1510 b, 1710 a, 1710 b thendetermines whether there are any more bits left in the bit stream. (Step1960) If there are, it returns to Step 1920 and breaks off another bbits. If the bit stream has ended, then the transmission process ends.(Step 1970)

As shown in FIG. 20, the receiver 1520, 1720 begins by receiving npulses transmitted by receiving a code word, comprising n pulses, fromthe transmitter 1510 a, 1510 b, 1710 a, 1710 b. (Step 2010)

If pseudo-random scrambling has been used on the received code word, thereceiver 1720 then multiplies the received code word by a predictablepseudo-random sequence. (Step 2020) In embodiments where nopseudo-random scrambling is used, or in portions of a packet where nopseudo-random scrambling is used, the receiver 1720 can omit this step.

The receiver 1520, 1720 correlates the received code word with kpossible code words to generate 1^(st) through k^(th) correlation values(Step 2030), and then compares the 1^(st) through k^(th) correlationvalues to determine the b-bit data sequence that the received code wordrepresents. (Step 2040)

The receiver 1520, 1720 then outputs this b-bit data sequence (Step2050) and determines whether there is another code word to receive.(Step 2060) If there is another code word, the receiver 1520, 1720returns to Step 2010 and receives the next code word. If there are nomore code words to receive, the reception process ends. (Step 2070)

Subrates

In alternate embodiments it is also possible to use subrate codes toincrease the E_(b)/N_(o) for the transmitted signals. In this case theMBOK codes (e.g., the ones shown in Table 3) are concatenated with anappropriate overlay code. In other words, a 1/n subrate will have noverlay code elements, each a 1 or a −1. In operation, n sequential codewords will be provided, each multiplied by the corresponding overlaycode element.

This effectively extends the length of a basic code word by 2 or 4,which makes the symbol longer, which gives more energy per symbol. Thiscan provide 3 dB or 6 dB, respectively, of additional E_(b)/N_(o), whichcan translate into greater range. The cost of this is a reduction inrate of data transfer.

For example, if code word 1 from Table 2 were used with the ½ overlaycode of Table 5, the code word would be repeated twice, once multipliedby 1, and once multiplied by −1 to obtain: −1 1 1 1 1 1 −1 −1 −1 1 1 −11 −1 −1 −1 −1 −1 1 1 1 −1 −1 1 as a subrate code. TABLE 5 SubrateOverlay Codes Subrate Factor Overlay Code ½ 1 −1 ¼ 1 1 −1 −1

Although 1 1 −1 −1 is shown as an overlay code for a subrate of ¼,alternate embodiments could use and overlay code of 1 −1 1 −1.

Carrier Offset

In embodiments that use segments of a continuously generated oscillatingsignal it is also possible to include a carrier offset to the code wordsused for multiple overlapping networks. In this case, a basic frequencyused for the oscillating signal (called a carrier frequency) is offsetfor each of the networks by a unique offset value. Thus, each networkwill have nearly the same carrier frequency for its pulses, but nonewill be identical.

Tables 6A and 6B show examples of carrier offset values as they are usedin preferred embodiments of the present invention. Table 6A shows anembodiment having seven overlapping networks, and is exemplary ofembodiments having an odd number of overlapping networks. Table 6B showsan embodiment having four overlapping networks, and is exemplary ofembodiments having an even number of overlapping networks. This carrieroffset can work for any sort of pulse, whether a monopulse, a section ofan oscillating signal, etc. TABLE 6A Carrier Offset Values for up toSeven Overlapping Networks Network Identifier Carrier Offset Value 0 −9MHz 1 −6 MHz 2 −3 MHz 3 Unchanged 4 +3 MHz 5 +6 MHz 6 +9 MHz

As shown in Table 6A, the carrier frequency (also called a centerfrequency) of each network is adjusted from the nominal carrierfrequency by the appropriate carrier offset value. When an odd number ofoverlapping networks are provided for, one may use the nominal carrierfrequency, while the remaining networks use an offset carrier frequency.Preferably the offset carrier frequencies are symmetrical around thenominal carrier frequency. TABLE 6B Carrier Offset Values for up to FourOverlapping Networks Network Identifier Carrier Offset Value 0 −9 MHz 1−3 MHz 2 +3 MHz 3 +9 MHz

As shown in Table 6B, the carrier frequency of each network is adjustedfrom the nominal carrier frequency by the appropriate carrier offsetvalue. When an even number of overlapping networks are provided for,none of the networks use the nominal carrier frequency. Instead eachnetwork uses an offset carrier frequency. Preferably the offset carrierfrequencies are symmetrical around the nominal carrier frequency.

Although Tables 6A and 6B show offset values for four and sevennetworks, more or fewer overlapping networks could be accommodated.Also, while in this embodiment the offset values are multiples of 3 MHz,in alternate embodiments the offset value could be changed. In someembodiments the offsets could use a different step value, or even haveno set step value at all, varying from each other according to no setpattern. The practical limit of the offset values can be used is thetuning range of the oscillator used.

In one preferred embodiment two separate bands are used, a high band anda low band. The high band has a nominal carrier frequency of 8.208 GHz,and the low band has a nominal carrier frequency of 4.104 GHz.

In operation, the selection of the carrier offset value used by a givennetwork will preferably be determined by the network's coordinatordevice during the initial scan prior to initiating the network. In thiscase, the network coordinator preferably selects a carrier offset valuethat is not in use by any other detected network in the area. Preferablythis will be done at the same time that the network coordinator choosesa code word set for the network. In fact, the codeword set and thecarrier offset will preferably be linked, each new network choosing alinked set to use.

As noted above, the use of the individual code words provides a degreeof channel separation between overlapping networks during preambleacquisition, limited only by the cross-correlation properties of thecode word set used by each network. The use of the carrier offset valuesupplements this separation by providing a degree of channel drift thatkeeps the channels used by each network from becoming stationary withrespect to the other channels.

This is helpful because although the code words limit the number ofconflicts between the signals of overlapping networks, they cannoteliminate them. If the center frequencies (i.e., carrier frequencies)used by each network were identical, then any conflicts between codes ofoverlapping networks would be fixed in time relative to each other.

However, if the two (or more) overlapping networks each have a slightlyoffset center frequency, the chipping phases of the networks will driftwith time. This means that any significant interference between any twonetworks will fade away with time as the chipping phases of each networkdrift with respect to each other. And while the differing centerfrequencies also means that any interferences will also come back, theirtransitory nature means that they can often be corrected for throughsignal processing, e.g., through the use of forward error correction(FEC).

Therefore, in embodiments using pulses formed from segments of anoscillating signal, the use of a carrier offset can reduce the chance ofcontinued interference between two overlapping networks, allowing anyinterference to be of limited duration and therefore potentiallycorrectable.

Forward Error Correction

In the preferred embodiments of the present invention, different typesof forward error correction (FEC) could be used. For example, the systemcould use convolutional FEC (e.g., at ½ rate, ⅔ rate and ¾ rate),Reed-Solomon FEC (255, 223), as well as Concatenated Convolutional &Reed-Solomon FEC.

Each of these varieties of FEC has unique characteristics that can beexploited depending upon the application. Convolutional FEC has lowlatency, limited speed, and moderate coding gain. Reed-Solomon FEC hasmoderate latency, high speed, and moderate coding gain. Concatenated FEChas high coding gain, high latency, and limited speed. As a result ofthese variations, different FEC methods can be used in differentsituations, as desired.

Clear Channel Assessment (CCA)

In order to operate more efficiently, it is desirable that a UWB networkbe able to determine quickly whether a given channel is being used byanother device or not. This is particularly useful in a carrier sensemultiple access (CSMA) environment, though it can be helpful in anyimplementation where quick scanning is desirable. The individualchannels that may be scanned are m-cycle BPSK channels (three-cycle BPSKin the disclosed embodiments) separated by frequency. This process canbe called carrier sense or clear channel assessment.

In previous implementations, a full acquisition process was required todetermine if a specific channel was clear (i.e., unused by anotherdevice) or not. However, an alternate approach allows for an assessmentof whether the channel is clear to be performed quickly.

FIGS. 21A and 21B are block diagrams showing circuits for performing arapid clear channel assessment according to preferred embodiments of thepresent invention. In preferred embodiments these circuits preferablyuse segments of an oscillating signal as pulses. However, alternateembodiments could use different pulse structures.

First Preferred Embodiment of CCA Circuitry

As shown in FIG. 21A, the clear channel assessment circuit 2100 aincludes an antenna 2105, an RF front end 2110, a first mixer 2120, asecond mixer 2125, a base oscillator 2130, a 0/90 phase shifter 2135, afirst low pass filter (LPF) 2140, a second LPF 2145, a first squaringcircuit 2150, a second squaring circuit 2155, an adder 2160, aninverting buffer 2165, a third mixer 2170, a doubling buffer 2175, athird LPF 2180, an absolute value circuit 2182, an automatic gaincontrol (AGC) loop filter 2184, and a decimated fast Fourier transform(FFT) 2185.

Although this circuit may be performed entirely using analog circuitryin some embodiments, in a preferred embodiment analog-to-digitalconverter (ADCs) can be used at some point along the signal stream toperform part of the operation digitally. In a preferred embodiment, afirst ADC 2190 is placed between the first LPF 2140 and the firstsquaring circuit 2150, and a second ADC 2195 is placed between thesecond LPF 2145 and the second squaring circuit 2155. However, inalternate embodiments, the number and placement of ADCs could bealtered, or they could be eliminated altogether.

In operation, the clear channel assessment circuit 2190 a operates asfollows. A signal comes in at the antenna 2105. This signal is sentthrough the front end 2110, which preferably includes a variable gainamplifier controlled by feedback from the AGC loop filter 2180. Once theincoming signal has been processed through the front end 2110, it isprovided to inputs in both the first and second mixers 2120 and 2125.These two mixers 2120 and 2125 mark the beginning of what can be calledI and Q paths for the incoming signal, and this process of breaking thesignal up into I and Q paths can be called I/Q demodulation.

The base oscillator 2130 provides a base oscillating signal at afrequency of F_(c). Preferably this base oscillating signal is asinusoidal signal, though alternate waveforms can be used in alternateembodiments. F_(c) is the center frequency of the particular bandwidthbeing used. In the preferred embodiment two bands are used, one centeredon 4.104 GHz and the other centered on 8.208 GHz. Thus, if the circuit2100 a is for the low band, F_(c) is 4.104 GHz, and if the circuit 2100a is for the high band, F_(c) is 8.208 GHz. This can be varied inalternate embodiments.

The base oscillating signal output from the base oscillator 2130 is sentthrough the 0/90 phase shifter, which outputs first and secondoscillating signals that are out of phase from each other by 90 degrees.The first oscillating signal is provided to an input of the first mixer2120, and the second oscillating signal is provided to an input of thesecond mixer 2125.

The phase difference between the first and second oscillating signalscan be accomplished by allowing one copy of the base oscillating signalto pass unchanged, while another copy is shifted 90 degrees. Otherembodiments could manipulate the base oscillating signal in other waysto provide the first and second oscillating signals. In the embodimentof FIG. 21A, the first oscillating signal is the same phase as the baseoscillating signal, while the second oscillating signal is delayed inphase by 90 degrees from the base oscillating signal. This could bealtered in alternate embodiments, so long as the first and secondoscillating signals were out of phase by 90 degrees.

The first mixer 2120 mixes the first oscillating signal and the signalreceived from the front end 2110 and provides a first mixed signal tothe first LPF 2140. Similarly, the second mixer 2125 mixes the secondoscillating signal and the signal received from the front end 2110 andprovides a second mixed signal to the second LPF 2145.

The first and second LPFs 2140 and 2145 are preferably root raise cosinefilters with a cutoff frequency proportional to the modulated signal, asis common for root raised cosine Nyquist filters. Other filter types andbandwidths can be used in alternate embodiments, however. In thepreferred embodiment using high and low bands, the cutoff frequency usedfor the low band is 684 MHz, and the cutoff frequency used for the highband is 1.368 GHz. This can be modified in alternate embodiments.

The output of the first LPF 2140 is provided to both the first squaringcircuit 2150 and the third mixer 2170, while the output of the secondLPF 2145 is provided to both the second squaring circuit 2155 and thethird mixer 2170.

The first squaring circuit 2150 squares the output of the first LPF 2140to provide a first squared signal, while the second squaring circuit2155 squares the output of the second LPF 2145 to provide a secondsquared signal.

The third mixer 2170 mixes the output of the first LPF 2140 and theoutput of the second LPF 2145 to provide a third mixed signal.

The inverting buffer 2165 inverts the second squared signal to providean inverted signal, while the doubling buffer doubles the third mixedsignal to provide a doubled signal.

The adder 2160 adds the first squared signal, the inverted signal, andthe doubled signal to produce an adder output signal.

One way to look at the clear channel assessment circuit 2100 a is toconsider that it breaks the incoming signal into a real portion x outputfrom the first LPF 2140, and an imaginary portion y output from thesecond LPF 2145. The square of the incoming signal can also becalculated as the square of the sum of the real and imaginary portionsof the incoming signal, as follows: $\begin{matrix}\begin{matrix}{{{Square}{\quad\quad}{of}{\quad\quad}{Incoming}\quad{Signal}} = ( {x + {jy}} )^{2}} \\{= {x^{2} + {j\quad 2{xy}} - y^{2}}} \\{= {( {x^{2} - y^{2}} ) + {j\quad 2{xy}}}}\end{matrix} & (18)\end{matrix}$

Thus, the output of the adder 2160 represents the real portion of thesquare of the input signal, while the output of the doubling bufferrepresents the imaginary portion of the input signal.

The third LPF 2180 serves to remove double frequency components in thesquared input signal. In the preferred embodiment the third LPF 2180 hasa cutoff frequency of 20 MHz.

The absolute value circuit 2182 takes the output of the third LPF 2180and gives it a positive magnitude.

The ACG loop filter 2180 is preferably a first order control loop filterwith an output proportional to the error signal at the input. Otherfilter types are possible in alternate embodiments, however. The ACGloop filter 2184 filters the output of the absolute value circuit 2182and provides the result to the front end 2120 as a feedback signal.

The output of the third LPF filter 2180 is also provided to thedecimated FFT circuitry 2185 as an input signal, which performs adecimated fast Fourier transform on the signal, moving the signal fromthe time domain to the frequency domain. The result of this decimatedfast Fourier transform is a clear channel assessment (CCA) signal thatindicates whether another network is on the air in the channel beinglistened to.

The current device compares the CCA against a noise baseline todetermine if another device is using the channel in question. If the CCAsignal is above a set threshold, then the device determines that thechannel being investigated is in use; if the CCA signal is not above thethreshold, then the device determines that the channel beinginvestigated is not in use. The noise baseline and associated thresholdscan be determined by observation of unused channels, or by other knownalgorithms.

In a preferred embodiment, the clear channel assessment circuit 2100 aoperates with analog circuitry up until the first and second LPFs 2140and 2145, and operates with digital circuitry thereafter. Therefore, inthis embodiment the first ADC 2190 is inserted between the first LPF2140 and the first squaring circuit 2150, and the second ADC 2195 isinserted between the second LPF 2145 and the second squaring circuit2155. In alternate embodiments the analog/digital line could be moved,or the whole operation could be performed in the analog realm.

This first preferred embodiment of the CCA circuitry requires that thebase oscillator 2130 be very accurate, which can require morecomplicated and expensive circuitry. Therefore, a second preferredembodiment is provided that allows for a feedback control of thefrequency of the base oscillator 2130.

Second Preferred Embodiment of CCA Circuitry

As shown in FIG. 21B, the clear channel assessment circuit 2100 bincludes an antenna 2105, an RF front end 2110, a first mixer 2120, asecond mixer 2125, a base oscillator 2130, a first 0/90 phase shifter2135, a loop filter 2132, a voltage-controlled oscillator (VCO) 2134, asecond 0/90 phase shifter 2136, a first low pass filter (LPF) 2140, asecond LPF 2145, a first squaring circuit 2150, a second squaringcircuit 2155, an adder 2160, an inverting buffer 2165, a third mixer2170, a fourth mixer 2172, a fifth mixer 2174, a doubling buffer 2175,an absolute value circuit 2182, an automatic gain control (AGC) loopfilter 2184, and a decimated fast Fourier transform (FFT) 2185. Elementsin FIG. 21B that have the same reference numbers as in FIG. 21A operatein a similar manner.

Although this circuit may be performed entirely using analog circuitryin some embodiments, in a preferred embodiment analog-to-digitalconverter (ADCs) can be used at some point along the signal stream toperform part of the operation digitally. In a preferred embodiment, afirst ADC 2190 is placed between the first LPF 2140 and the firstsquaring circuit 2172, and a second ADC 2195 is placed between thesecond LPF 2145 and the second squaring circuit 2174. However, inalternate embodiments, the number and placement ADCs could be altered,or they could be eliminated altogether.

In operation, the clear channel assessment circuit 2100 b operates asfollows. A signal comes in at the antenna 2105. This signal is sentthrough the front end 2110, which preferably includes a variable gainamplifier controlled by feedback from the AGC loop filter 2180. Once theincoming signal has been processed through the front end 2110, it isprovided to inputs in both the first and second mixers 2120 and 2125.These two mixers 2120 and 2125 mark the beginning of what can be calledI and Q paths for the incoming signal, and this process of breaking thesignal up into I and Q paths can be called I/Q demodulation.

The base oscillator 2130 provides a base oscillating signal at afrequency of F_(c). Preferably this base oscillating signal is asinusoidal signal, though alternate waveforms can be used in alternateembodiments. F_(c) is the center frequency of the particular bandwidthbeing used. In the preferred embodiment two bands are used, one centeredon 4.104 GHz and the other centered on 8.208 GHz. Thus, if the circuit2100 a is for the low band, F_(c) is 4.104 GHz, and if the circuit 2100b is for the high band, F_(c) is 8.208 GHz. This can be varied inalternate embodiments.

The base oscillating signal output from the base oscillator 2130 is sentthrough the first 0/90 phase shifter 2135, which outputs first andsecond oscillating signals that are out of phase from each other by 90degrees. The first oscillating signal is provided to an input of thefirst mixer 2120, and the second oscillating signal is provided to aninput of the second mixer 2125.

The first 0/90 phase shifter 2135 can achieve the phase differencebetween the first and second oscillating signals by allowing one copy ofthe base oscillating signal to pass unchanged, while another copy isshifted 90 degrees. However, other embodiments could manipulate the baseoscillating signal in other ways to provide the first and secondoscillating signals. In the embodiment of FIG. 21B, the firstoscillating signal is the same phase as the base oscillating signal,while the second oscillating signal is delayed in phase by 90 degreesfrom the base oscillating signal. This could be altered in alternateembodiments, so long as the first and second oscillating signals wereout of phase by 90 degrees.

The first mixer 2120 mixes the first oscillating signal and the signalreceived from the front end 2110 and provides a first mixed signal tothe first LPF 2140. Similarly, the second mixer 2125 mixes the secondoscillating signal and the signal received from the front end 2110 andprovides a second mixed signal to the second LPF 2145.

The first and second LPFs 2140 and 2145 are preferably root raise cosinefilters with a cutoff frequency proportional to the modulated signal, asis common for root raised cosine Nyquist filtering. Other filter typesand bandwidth can be used in alternate embodiments, however. In thepreferred embodiment using high and low bands, the cutoff frequency usedfor the low band is 684 MHz, and the cutoff frequency used for the highband is 1.368 GHz. This can be modified in alternate embodiments.

The fourth mixer 2172 receives the output of the first LPF 2140 and athird oscillating signal received from the second 0/90 phase shifter2136, and mixes the two to provide a fourth mixed signal. The fifthmixer 2174 receives the output of the second LPF 2145 and a fourthoscillating signal received from the second 0/90 phase shifter 2136, andmixes the two to provide a fifth mixed signal.

The first squaring circuit 2150 squares the fourth mixed signal toprovide a first squared signal, while the second squaring circuit 2155squares the fifth mixed signal to provide a second squared signal.

The third mixer 2170 mixes the fourth and fifth mixed signals to providea third mixed signal.

The loop filter 2132 is preferably a type 2 second order lead-lag loopfilter that serves to integrate the error signal from the third mixer2170, controlling the VCO 2134.

The output of the loop filter 2132 is then used to control the frequencyof the VCO 2134, which produces a corrective oscillating signal. Thiscorrective oscillating signal is used to correct the frequency errorintroduced by the base oscillating signal produced by the baseoscillator 2130.

Preferably the VCO 2134 has a frequency that is in the range of about 0MHz to 10 MHz, depending upon the output of the loop filter 2132.

The corrective oscillating signal output from the VCO 2134 is sentthrough the second 0/90 phase shifter 2136, which outputs third andfourth oscillating signals that are out of phase from each other by 90degrees. The third oscillating signal is provided to an input of thefourth mixer 2172, and the fourth oscillating signal is provided to aninput of the fifth mixer 2174.

The second 0/90 phase shifter 2136 can achieve the phase differencebetween the third and fourth oscillating signals by allowing one copy ofthe corrective oscillating signal to pass unchanged, while another copyis shifted 90 degrees. However, other embodiments could manipulate thecorrective oscillating signal in other ways to provide the third andfourth oscillating signals. In the embodiment of FIG. 21B, the thirdoscillating signal is the same phase as the corrective oscillatingsignal, while the fourth oscillating signal is delayed in phase by 90degrees from the corrective oscillating signal. This could be altered inalternate embodiments, so long as the third and fourth oscillatingsignals were out of phase by 90 degrees. The relative phases of thethird and fourth oscillating with respect to the first and secondoscillating signals is unimportant.

The inverting buffer 2165 inverts the second squared signal to providean inverted signal, while the doubling buffer doubles the third mixedsignal to provide a doubled signal.

The adder 2160 adds the first squared signal, the inverted signal, andthe doubled signal to produce an adder output signal.

As noted above, one way to look at the clear channel assessment circuit2100 b is to consider that it breaks the incoming signal into a realportion x output from the first LPF 2140, and an imaginary portion youtput from the second LPF 2145. And based on Equation (18), the outputof the adder 2160 represents the real portion of the square of the inputsignal, while the output of the doubling buffer represents the imaginaryportion of the input signal.

The third LPF 2180 serves to remove double frequency components in thesquared input signal. In the preferred embodiment the third LPF 2180 hasa cutoff frequency of 20 MHz.

The absolute value circuit 2182 takes the output of the third LPF 2180and gives it a positive magnitude.

The ACG loop filter 2180 is preferably a first order control loop filterwith an output proportional to the error signal at the input. Otherfilters are possible, however, in alternate embodiments. The ACG loopfilter 2184 filters the output of the absolute value circuit 2182 andprovides the result to the front end 2110 as a feedback signal.

The output of the third LPF filter 2180 is also provided to thedecimated FFT circuitry 2185 as an input signal, which performs adecimated fast Fourier transform on the signal, moving the signal fromthe time domain to the frequency domain. The result of this decimatedfast Fourier transform is a clear channel assessment (CCA) signal thatindicates whether another network is on the air in the channel beinglistened to.

The current device compares the CCA against a noise baseline todetermine if another device is using the channel in question. If the CCAsignal is above a set threshold, then the device determines that thechannel being investigated is in use; if the CCA signal is not above thethreshold, then the device determines that the channel beinginvestigated is not in use. The noise baseline and associated thresholdscan be determined by observation of unused channels, or by other knownalgorithms.

In a preferred embodiment, the clear channel assessment circuit 2100 boperates with analog circuitry up until the first and second LPFs 2140and 2145, and operates with digital circuitry thereafter. Therefore, inthis embodiment the first ADC 2190 is inserted between the first LPF2130 and the first squaring circuit 2172, and the second ADC 2195 isinserted between the second LPF 2135 and the second squaring circuit2174. In alternate embodiments the analog/digital line could be moved,or the whole operation could be performed in the analog realm.

Choosing Frequency Offsets

Using chipping rate offsets between networks forces RMScross-correlation conditions between network code words. Because ofthis, there is a required minimum frequency offset in order to insurethat cross-correlation errors do not cause burst errors. For offsetsless than the minimum, cross-correlation spikes can cause burst errors,which will require some sort of FEC capable of dealing with bursterrors.

This makes it necessary to balance the increased costs of FEC operationsto address burst errors with the costs of providing hardware complexenough to use frequency offsets adequate to guarantee single errors.

The following analysis will address the question of how much of afrequency offset is required between two networks to avoid burst errors.

Assuming that the symbol (code word) duration from two different sourceswill differ by an amount τ; that is:T _(S2) =T _(S1)+τ,  (19)

the chipping rates of the first and second sources can be described asfollows: $\begin{matrix}{f_{C\quad 1} = \frac{N}{T_{S\quad 1}}} & (20) \\{f_{C\quad 2} = {\frac{N}{T_{S\quad 2}} = {\frac{N}{T_{S\quad 1} + \tau} = {\frac{N}{T_{S\quad 1}}{\{ \frac{1}{1 + \frac{\tau}{T_{S\quad 1}}} \}.}}}}} & (21)\end{matrix}$

where f_(C1) is the chipping rate for the first source, f_(C2) is thechipping rate for the second source, N is the code word length, T_(S1)is the symbol duration of a first source, T_(S2) is the symbol durationof a second source, and τ is the difference in symbol duration betweenthe first and second sources.

Equation 21 can be expanded using a binomial series, and can betruncated to yield: $\begin{matrix}{\begin{matrix}{f_{C\quad 2} = {\frac{N}{T_{S\quad 1}}\lbrack {1 - \frac{\tau}{T_{S\quad 1}}} \rbrack}} \\{= {f_{C\quad 1}\lbrack {1 - \frac{\tau}{T_{S\quad 1}}} \rbrack}} \\{= {f_{C\quad 1} - {f_{C\quad 1} \cdot \frac{\tau}{T_{S\quad 1}}}}}\end{matrix}.} & (22)\end{matrix}$

The offset frequency difference Δf can then be determined as:$\begin{matrix}{\begin{matrix}{{\Delta\quad f} = {f_{C\quad 1} - f_{C\quad 2}}} \\{= {f_{C\quad 1} - ( {f_{C\quad 1} - {f_{C\quad 1} \cdot \frac{\tau}{T_{S\quad 1}}}} )}} \\{= {f_{C\quad 1} - f_{C\quad 1} + {f_{C\quad 1} \cdot \frac{\tau}{T_{S\quad 1}}}}} \\{= {f_{C\quad 1} \cdot \frac{\tau}{T_{S\quad 1}}}} \\{= {f_{C\quad 1} \cdot \frac{\tau}{N/f_{C\quad 1}}}} \\{= {f_{C\quad 1}^{2}\frac{\tau}{N}}}\end{matrix}.} & (23)\end{matrix}$

The values of N and f_(C2) are determined by the network. The value of τcan be determined by the “time width” of the cross-correlation, which inturn is determined by the wavelet autocorrelation. It is thereforepossible to determine what the required frequency difference Δf (i.e.,the required offset frequency) is to decorrelate between two symbols andavoid burst errors.

In one preferred embodiment, N=24 (i.e., the code word is 24 chipslong), and f_(C1)=2.736 Gcps (i.e., the chipping rate is 2.736 Gcps),and the autocorrelation 3 dB time width for 70 pS peak-to-peak waveletsis ±10 pS.

Using equation 23, this results in a the following minimum frequencydifference to decorrelate between two symbols: $\begin{matrix}{\begin{matrix}{{\Delta\quad f} = {\frac{10 \times 10^{- 12}}{24}( {2.736 \times 10^{9}} )^{2}}} \\{= {{3.119\quad{MHz}} \approx {3.12\quad{MHz}}}}\end{matrix}.} & (24)\end{matrix}$

In other words, the frequency offset between chips must be about 3.12MHz to decorrelate between two symbols. If the frequency offset ischosen to be 3.12 MHz or greater, then no burst errors will occur. Ifthe frequency offset is chosen to be below 3.12 MHz, then burst errorswill occur.

The length of burst errors will be determined by the chosen frequencyoffset. Using the parameters from Equation 23, consider if the offsetfrequency Δf were only 1 MHz. The difference in symbol duration betweenthe first and second sources τ can be calculated as: $\begin{matrix}{\tau = {\frac{{N \times \Delta}\quad f}{f_{C}^{2}} = {\frac{24 \times 10^{6}}{( {2.736 \times 10^{9}} )^{2}} = {3.2{pS}}}}} & (25)\end{matrix}$

The error burst length can then be considered to be approximately theautocorrelation time width divided by difference in symbol durationbetween the first and second sources τ: $\begin{matrix}{{{Error}\quad{Burst}{\quad\quad}{Length}\text{:}} \approx \frac{20{pS}}{3.2{pS}} \approx 6} & (26)\end{matrix}$

The error burst length shows how long the error condition will persistbefore the offending code words drift apart in phase. This is determinedby dividing the possible variation of the autocorrelation 3 dB timewidth (20 pS, given the ±10 pS range) by τ.

As noted above with respect to Table 6B, in a preferred embodiment thefrequency offsets for the chips are −9 MHz, −3 MHz, +3 MHz, and +9 MHz,which correspond to offsets of −3 MHz, −1 MHz, +1 MHz, and +3 MHz forthe clock used to form the chips. This means that for two channels rightnext to each other, the frequency offset Δf will be 2 MHz, which gives adifference in symbol duration τ between devices on the two channels as:$\begin{matrix}{\tau = {\frac{{N \times \Delta}\quad f}{f_{C}^{2}} = {\frac{24 \times 2 \times 10^{6}}{( {2.736 \times 10^{9}} )^{2}} = {6.4{pS}}}}} & (27)\end{matrix}$

with an error burst length of: $\begin{matrix}{{{{Error}\quad{Burst}\quad{Length}\text{:}} \approx \frac{20\quad{pS}}{6.4\quad{pS}} \approx 3},} & (28)\end{matrix}$

which can be corrected through the use of FEC.

However, for two channels that are not adjacent, the frequency offset Δfwill be 4 MHz or 6 MHz, which eliminates the risk of burst errors.

Therefore this implementation either eliminates burst errors or allowsburst errors at a rate that can be addressed through the use of FEC. Inthe preferred embodiment, the coordinator of a network will preferablyassign channels (i.e., frequency offsets) in such a way as to maximizethe frequency offsets between the devices. In this way, the use of FECwill be used only when it cannot be avoided.

Clear Channel Determination

A multiple network environment can be modeled as a vector of signalsV _(S)(t)=[S ⁻³(t) S ⁻²(t) S ⁻¹(t) S ₀(t) S ₊₁(t) S ₊₂(t) S ₊₃(t)]

where S_(i)(t)=m_(i)(t)*cos{(ω₀+ω_(i))t}, ω_(i) is the frequency offset,and m_(i) is the time dependent modulation. This vector will beprocessed by a square law device (e.g., the squaring circuits 2150 and2155 in FIGS. 21A and 21B).

The matrix product is given as of this squaring function is:$\begin{matrix}{{{V_{S}^{T}(t)}*{V_{S}(t)}} = \begin{matrix}{S_{- 3}^{2}(t)} & {{S_{- 3}(t)}{S_{- 2}(t)}} & {{S_{- 3}(t)}{S_{- 1}(t)}} & {{S_{- 3}(t)}{S_{0}(t)}} & {{S_{- 3}(t)}{S_{+ 1}(t)}} & {{S_{- 3}(t)}{S_{+ 2}(t)}} & {{S_{- 3}(t)}{S_{+ 3}(t)}} \\{{S_{- 2}(t)}S_{-}3(t)} & {S_{- 2}^{2}(t)} & {{S_{- 2}(t)}{S_{- 1}(t)}} & {{S_{- 2}(t)}{S_{0}(t)}} & {{S_{- 2}(t)}{S_{+ 1}(t)}} & {{S_{- 2}(t)}{S_{+ 2}(t)}} & {{S_{- 2}(t)}{S_{+ 3}(t)}} \\{{S_{- 1}(t)}{S_{- 3}(t)}} & {{S_{- 1}(t)}{S_{- 2}(t)}} & {S_{- 1}^{2}(t)} & {{S_{- 1}(t)}{S_{0}(t)}} & {{S_{- 1}(t)}{S_{+ 1}(t)}} & {{S_{- 1}(t)}{S_{+ 2}(t)}} & {{S_{- 1}(t)}{S_{+ 3}(t)}} \\{{S_{0}(t)}{S_{- 3}(t)}} & {{S_{0}(t)}{S_{- 2}(t)}} & {{S_{- 0}(t)}{S_{- 1}(t)}} & {S_{0}^{2}(t)} & {{S_{0}(t)}{S_{+ 1}(t)}} & {{S_{0}(t)}{S_{+ 2}(t)}} & {{S_{0}(t)}{S_{+ 3}(t)}} \\{{S_{+ 1}(t)}{S_{- 3}(t)}} & {{S_{+ 1}(t)}{S_{- 2}(t)}} & {{S_{+ 1}(t)}{S_{- 1}(t)}} & {{S_{+ 1}(t)}{S_{0}(t)}} & {S_{+ 1}^{2}(t)} & {{S_{+ 1}(t)}{S_{+ 2}(t)}} & {{S_{+ 1}(t)}{S_{+ 3}(t)}} \\{{S_{+ 2}(t)}{S_{- 3}(t)}} & {{S_{+ 2}(t)}{S_{- 2}(t)}} & {{S_{+ 2}(t)}{S_{- 1}(t)}} & {{S_{+ 2}(t)}{S_{0}(t)}} & {{S_{+ 2}(t)}{S_{+ 1}(t)}} & {S_{+ 2}^{2}(t)} & {{S_{+ 2}(t)}{S_{+ 3}(t)}} \\{{S_{+ 2}(t)}{S_{-}(t)}} & {{S_{+ 3}(t)}{S_{- 2}(t)}} & {{S_{+ 3}(t)}{S_{- 1}(t)}} & {{S_{+ 3}(t)}{S_{0}(t)}} & {{S_{+ 3}(t)}{S_{+ 1}(t)}} & {{S_{+ 3}(t)}{S_{+ 2}(t)}} & {S_{+ 3}^{2}(t)}\end{matrix}} & (29)\end{matrix}$

All the signals off the main diagonal represent the product of twouncorrelated spread spectrum signals which yields just another spreadspectrum signal (represents an increase in the noise floor). However,the trace represents the square-law product sum of the signals S_(i)²(t)=m_(i) ²(t)*cos²{(ω₀+ω_(i))t}. The expectation of each doublefrequency term is given by $\begin{matrix}{\overset{\_}{S_{i}^{2}(t)} = {\frac{1}{2}*\overset{\_}{m_{i}^{2}(t)}*\cos\{ {2( {\omega_{0} + \omega_{i}} )t} \}}} & (30)\end{matrix}$

where {overscore (m_(i) ²(t))}≈1. This shows that the trace termscollapse to a double frequency component and the cross-product terms(off main diagonal terms) simply raise the noise floor. Assuming eachnetwork uses a unique chipping rate offset, the output of the squaringloop can be used for network identification.

The output of the squarer in FIG. 21A is a spectral comb representingthe above trace terms with the noise floor set by the cross-productterms. This output is determined by the combination of the real andimaginary signal portions output from the third LPF 2180, and providedto the decimate FFT 2185.

FIGS. 23A and 23B are FFT graphs of a simulation of the output of thethird LPF 2180 of FIG. 21A with 3 terms and 7 terms (i.e., channels),respectively, according to a preferred embodiment of the presentinvention. In each simulation, the input signals were the same strengthand the frequency offset was 1 MHz, which corresponds to a squaredspectral line separation of 2 MHz.

In operation of the clear channel assessment circuit 2100 a, thedecimated FFT 2185 would perform the same FFT analysis of the signalsfrom the third LPF 2180 as shown in FIGS. 23A and 23B, except over asmaller bandwidth corresponding to the possible positions of spikes in agiven channel, and producing a single value as a result.

If the result output from the decimated FFT 2185 is above a determinedthreshold, then the device determines that there is another devicetransmitting over the given channel. If, however, the result output fromthe decimated FFT 2185 is below the threshold, then the devicedetermines that the given channel is unused.

As shown in FIGS. 23A and 23B, each graph has a noise floor with anumber of spikes corresponding to the number of terms used. And althoughthe signal strengths were the same in both simulations, the noise floorin FIG. 23B is roughly 10 dB higher than the noise floor of FIG. 23A.

Signal Acquisition

Signal acquisition in the disclosed UWB systems is preferably performedby code wheel spinning using a correlator 1560 as shown in FIGS.15A-17B. Once the timing of the chips (i.e., pulses) is determined, asliding correlator (e.g., the correlator 1560 from FIGS. 15A-15C)rotates through a full cycle of phases for the acquisition codeword todetermine where the symbol boundaries lie. Proper codeword phasing, andhence symbol alignment, is associated with the peak response of thecorrelator.

The present system provides several advantages over existing UWBsystems. As noted above, the current system performs a clear channeldetermination very quickly to allow more efficient use of a CSMAenvironment.

In addition, the present system allows three different preamble types,which can be used for different situations. By allowing the devices tochoose between each of these preambles, the system improves the abilityof the devices to provide for the shortest acquisition and maximum datarate allowed by the current transmission conditions.

A preferred embodiment of the present invention provides three differentpreambles, each with slightly different parameters. A short preamble isused for situations where a transmission is strong or otherwise requireslittle in the way of processing to pass between devices successfully; along preamble is used for situations where the transmission is weak andrequires maximum processing to pass successfully between devices; and anormal preamble is used for situations between these two extremes.Although three preamble types are used in the preferred embodiment,alternate embodiments could use more or fewer preamble choices, asdesired.

FIGS. 22A-22C are block diagrams of a short preamble, a normal preamble,and a long preamble, respectively, according to preferred embodiments ofthe present invention.

Short Preamble

As shown in FIG. 22A, a short preamble 2201 includes an automatic gaincontrol (AGC)/clear channel assessment (CCA) portion 2210, a shortsynchronization portion 2221, a start frame delimiter (SFD) 2230, and aphysical layer (PHY) header 2240.

The AGC/CCA portion 2210 is preferably a period of time set aside forCCA and AGC setting. The length of the AGC/CCA portion 2210 is set toprovide sufficient time for CCA and AGC. In the preferred embodimentdisclosed in FIG. 22B, the AGC/CCA portion 2210 is 4 μs long.

The short synchronization portion 2221 provides the receiver with aknown set of data that allows the receiving device to lock onto thechipping clock timing and the symbol clock timing of the transmitter. Inother words, it allows the receiver to synchronize with the phase of thepulses and symbols being transmitted. In the preferred embodimentdisclosed in FIG. 22A, the short synchronization portion 2221 is 5 μslong.

The SFD 2230 serves as a delimiter, indicating when the PHY header 2240starts. In the preferred embodiment disclosed in FIG. 22A, the SFD 2230is ½ μs long.

The PHY header 2240 provides time and information to allow the receivingdevice to perform what acquisition operations are necessary based on theparticular PHY layer being used. The specific parameters of the PHYheader 2240 will vary as different PHY layers are used.

The short preamble is useful for situations where a strong signal isreceived and little processing beyond a set minimum is required tosuccessfully acquire a signal. This includes signals being transmittedat short range (e.g., about one meter), situations where no rake isrequired, and situations where no decision feedback equalization (DFE)is required. In the preferred embodiment, this would give the shortestacquisition time, and would allow the maximum data transmission rate.

Normal Preamble

As shown in FIG. 22B, a normal preamble 2202 includes an AGC/CCA portion2210, a normal synchronization portion 2222, a normal decision feedbackequalization (DFE) training portion 2252, an SFD 2230, and a PHY header2240.

The AGC/CCA portion 2210 is preferably a period of time set aside forCCA and AGC setting. The length of the AGC/CCA portion 2210 is set toprovide sufficient time for CCA and AGC. In the preferred embodimentdisclosed in FIG. 22B, the AGC/CCA portion 2210 is 4 μs long.

The normal synchronization portion 2222 provides the receiver with aknown set of data that allows the receiving device to lock onto thechipping clock timing and the symbol clock timing of the transmitter. Inother words, it allows the receiver to synchronize with the phase of thepulses and symbols being transmitted. In the preferred embodimentdisclosed in FIG. 22A, the normal synchronization portion 2221 is 8 μslong.

The normal DFE training portion 2252 provides time and information forthe receiver to perform decision feedback equalization on the incomingsignal. In the preferred embodiment disclosed in FIG. 22B, the DFEtraining portion 2250 is 4 μs long.

The SFD 223.0 serves as a delimiter, indicating when the PHY header 2240starts. In the preferred embodiment disclosed in FIG. 22A, the SFD 2230is ½ μs long.

The PHY header 2240 provides time and information to allow the receivingdevice to perform what acquisition operations are necessary based on theparticular PHY layer being used. The specific parameters of the PHYheader 2240 will vary as different PHY layers are used.

The normal preamble is useful for ranges longer than minimal ranges(e.g., above one meter), cases where rake or DFE is required, or anycases where some amount of signal processing between a minimum andmaximum amount is needed. In the preferred embodiment, the normalpreamble would give a middling acquisition time, and would allow anaverage relative data transmission rate.

Long Preamble

As shown in FIG. 22C, a long preamble 2203 includes an AGC/CCA portion2210, a long synchronization portion 2223, a DFE training portion 2253,an SFD 2230, and a PHY header 2240.

The AGC/CCA portion 2210 is preferably a period of time set aside forCCA and AGC setting. The length of the AGC/CCA portion 2210 is set toprovide sufficient time for CCA and AGC. In the preferred embodimentdisclosed in FIG. 22B, the AGC/CCA portion 2210 is 4 μs long.

The long synchronization portion 2223 provides the receiver with a knownset of data that allows the receiving device to lock onto the chippingclock and the symbol clock of the transmitter. In other words, it allowsthe receiver to synchronize with the phase of the pulses beingtransmitted as well as the phase of the symbols being transmitted. Inthe preferred embodiment disclosed in FIG. 22C, the long synchronizationportion 2223 is 91.5 μs long.

The DFE training portion 2253 provides time and information for thereceiver to perform decision feedback equalization on the incomingsignal. In the preferred embodiment disclosed in FIG. 22C, the DFEtraining portion 2260 is 4 μs long.

The SFD 2230 serves as a delimiter, indicating when the PHY header 2240starts. In the preferred embodiment disclosed in FIG. 22A, the SFD 2230is ½ μs long.

The PHY header 2240 provides time and information to allow the receivingdevice to perform what acquisition operations are necessary based on theparticular PHY layer being used. The specific parameters of the PHYheader 2240 will vary as different PHY layers are used.

The long preamble is useful for long ranges (e.g., near maximum range),cases where rake or DFE are required, situations with antenna diversity(i.e., when more than one antenna is used, and it is necessary todetermine which of the plurality of antennas gives the strongest signaland should be used for the duration of the packet), and any casesmaximum signal processing is needed. In the preferred embodiment, thenormal preamble would require the maximum acquisition time, and wouldallow a minimum data transmission rate.

Determining Preamble

As shown above, the lengths of the synchronization portions 2221, 2222,and 2223 vary among the three preambles 2201, 2202, and 2203. Inpreferred embodiments, the short synchronization portion 2221 is shorterthan the normal synchronization portion 2222, which is shorter than thelong synchronization portion 2223.

In situations where a transmission has very good signal strength, thedevices can choose the short preamble 2201 to maximize transmissionspeed. In this case, acquisition time is minimized because the highsignal strength means that acquisition can be performed more quickly.And the shortened acquisition time means a greater data transmissionrate

In situations where a transmission has poor signal strength, the devicescan choose the long preamble 2203 to maximize transmission speed. Inthis case, acquisition time is lengthened at the expense of datatransmission rate. But because the signal is poor, greater time isneeded to achieve a successful acquisition.

In situations where the signal strength is neither very good nor poor,the devices can choose a normal preamble 2202 to balance acquisitiontime and transmission speed. In this case, an average acquisition timeis provided, which allows an average data transmission rate.

One way to determine the quality of the signal strength is to measurehow many requests for packet retransmission are required for any givenunit time. When the number of packet retransmission requests from thereceiving device is low, the signal is determined to be strong. When thenumber of packet retransmission requests from the receiving device ishigh, the signal is determined to be low. By setting thresholds for thenumbers of packet retransmission appropriately, the transitions betweenthe preambles can be arranged to meet the required quality of serviceneeded for the current network environment.

It is important, however, that there be some way for all devices to knowbefore any transmission what the preamble size will be. In a preferredembodiment, one preamble is set as a default. Any newly formed networkwill always start using the normal preamble 2202. Then, oncecommunication is established, the devices in the network can determine(by whatever means is provided in the network) whether to change to adifferent preamble.

For example, if after a time the data signal was found to be verystrong, the network might move from the normal preamble 2202 to theshort preamble 2201 in order to increase data transmission rate. Then,if for some reason the signal strength degrades and acquisition becomesharder, the network can return to the normal preamble 2202, or even moveto the long preamble 2203.

Regardless, by providing multiple preambles, the system allows thenetwork to adjust its data transmission rate based on the acquisitionrequirements imposed by the current signal strength.

CONCLUSION

Obviously, numerous modifications and variations of the presentinvention are possible in light of the above teachings. It is thereforeto be understood that within the scope of the appended claims, theinvention may be practiced otherwise than as specifically describedherein.

1. A method of acquiring incoming signals in a wireless network device,comprising: acquiring the incoming signals using a default acquisitionpreamble having a default synchronization period; evaluating theincoming signals to determine whether first signal parameters are met;changing to a first alternate acquisition preamble having a firstalternate synchronization period if the first signal parameters are met;evaluating the incoming signals to determine whether second signalparameters are met; and changing to a second alternate acquisitionpreamble having a second alternate synchronization period if the secondparameters are met, wherein the default synchronization period, thefirst alternate synchronization period, and the second alternatesynchronization period all have different values.
 2. A method ofacquiring incoming signals in a wireless network device, as recited inclaim 1, wherein the default acquisition preamble is a normal preamblehaving a normal synchronization period, wherein the first acquisitionalternate preamble is a short preamble having a short synchronizationperiod, wherein the second alternate acquisition preamble is a longpreamble having a long synchronization period, and wherein the normalsynchronization period is longer than the short synchronization period,and the long synchronization period is longer than the normalsynchronization period.
 3. A method of acquiring incoming signals in awireless network device, as recited in claim 2, wherein the first signalparameters are met if the signal strength is above a first threshold,and wherein the second signal parameters are met if the signal strengthis below a second threshold.
 4. A method of acquiring incoming signalsin a wireless network device, as recited in claim 2, wherein the firstsignal parameters are met if a number of requested packetretransmissions per unit time in the wireless network device is below afirst threshold, and wherein the second signal parameters are met if thenumber of requested packet retransmissions per unit time in the wirelessnetwork device is above a second threshold.
 5. A method of acquiringincoming signals in a wireless network device, as recited in claim 2,wherein the normal acquisition preamble includes a normal decisionfeedback equalization training period, wherein the long acquisitionpreamble includes a long decision feedback equalization training period,and wherein the long decision feedback equalization training period islonger than the normal decision feedback equalization training period.6. A method of acquiring incoming signals in a wireless network device,as recited in claim 1, wherein the default acquisition preamble is ashort preamble having a short synchronization period, wherein the firstacquisition alternate preamble is a normal preamble having a normalsynchronization period, wherein the second alternate acquisitionpreamble is a long preamble having a long synchronization period, andwherein the normal synchronization period is longer than the shortsynchronization period, and the long synchronization period is longerthan the normal synchronization period.
 7. A method of acquiringincoming signals in a wireless network device, as recited in claim 6,wherein the first signal parameters are met if the signal strength isbelow a first threshold but not below a second threshold, and whereinthe second signal parameters are met if the signal strength is belowboth the first and second thresholds.
 8. A method of acquiring incomingsignals in a wireless network device, as recited in claim 6, wherein thefirst signal parameters are met if a number of requested packetretransmissions per unit time in the wireless network device is above afirst threshold but not above a second threshold, and wherein the secondsignal parameters are met if the number of requested packetretransmissions per unit time in the wireless network device is aboveboth the first and second thresholds.
 9. A method of acquiring incomingsignals in a wireless network device, as recited in claim 6, wherein thenormal acquisition preamble includes a normal decision feedbackequalization training period, wherein the long acquisition preambleincludes a long decision feedback equalization training period, andwherein the long decision feedback equalization training period islonger than the normal decision feedback equalization training period.10. A method of acquiring incoming signals in a wireless network device,as recited in claim 1, wherein the default acquisition preamble is along preamble having a long synchronization period, wherein the firstacquisition alternate preamble is a normal preamble having a normalsynchronization period, wherein the second alternate acquisitionpreamble is a short preamble having a short synchronization period, andwherein the normal synchronization period is longer than the shortsynchronization period, and the long synchronization period is longerthan the normal synchronization period.
 11. A method of acquiringincoming signals in a wireless network device, as recited in claim 10,wherein the first signal parameters are met if the signal strength isabove a first threshold but not above a second threshold, and whereinthe second signal parameters are met if the signal strength is aboveboth the first and second thresholds.
 12. A method of acquiring incomingsignals in a wireless network device, as recited in claim 10, whereinthe first signal parameters are met if a number of requested packetretransmissions per unit time in the wireless network device is below afirst threshold but not below a second threshold, and wherein the secondsignal parameters are met if the number of requested packetretransmissions per unit time in the wireless network device is belowboth the first and second thresholds.
 13. A method of acquiring incomingsignals in a wireless network device, as recited in claim 10, whereinthe normal acquisition preamble includes a normal decision feedbackequalization training period, wherein the long acquisition preambleincludes a long decision feedback equalization training period, andwherein the long decision feedback equalization training period islonger than the normal decision feedback equalization training period.14. A method of acquiring incoming signals in a wireless network device,as recited in claim 1, wherein the wireless network device is anultrawide bandwidth device.
 15. A method of acquiring incoming signalsin a wireless network device, comprising: acquiring the incoming signalsusing a default acquisition preamble having a default synchronizationperiod; evaluating the incoming signals to determine whether i^(th)signal parameters are met; and changing to an i^(th) alternateacquisition preamble having an i^(th) synchronization period if thei^(th) signal parameters are met, wherein k is an integer greater than1, wherein i is an integer that varies from 1 to k, wherein the firstthrough k^(th) synchronization periods all have different values, andwherein the first through k^(th) signal parameters are mutuallyexclusive.
 16. A method of acquiring incoming signals in a wirelessnetwork device, as recited in claim 15, wherein the wireless networkdevice is an ultrawide bandwidth device.